IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 67, NO. 3, MARCH 2019 1957 Communication Quarter-Wave Balun Fed Vivaldi Antenna Pair for V2X Communication Measurement Philip Ayiku Dzagbletey , Jae-Yeon Shim , and Jae-Young Chung Abstract— A dual-polarized Vivaldi antenna pair has been designed for use in a vehicle-to-anything (V2X) measurement facility. The operating bandwidth is from 560 MHz to 7.7 GHz (173% fractional bandwidth), covering all the existing V2X communication frequencies. The ultrawideband (UWB) property was achieved by integrating quarter-wave baluns at the feeding point of two orthogonally crossed Vivaldi antennas. The measured port isolation between the two antennas is better than −28 dB. The cross-polarization discrimination of better than 17.2 dB and antenna gain of up to 9.2 dB have also been realized across the whole bandwidth. Three of the designed antennas were mounted on a vehicle testing facility and were used to measure the radiation patterns of a roof-top shark antenna under four different V2X scenarios. That is, vehicle-to-pedestrian, vehicle-to-vehicle, and vehicle-to-infrastructure and vehicle-to-network. The results have also been presented in this communication. Index Terms— Antenna measurement, balun, dual polarization, quarter-wave tapering, tapered slot antenna (TSA), vehicle-to-anything (V2X) communication, Vivaldi antenna. Fig. 1. models. V2X Measurement facility with three mounted Vivaldi antenna I. I NTRODUCTION Vehicle-to-anything (V2X) communication is one of the key technologies to realize the intelligent transport system. Car makers and broadband network service providers are rigorously discussing the implementations of V2X systems based on two strong candidates of wireless communication standards: the IEEE 802.11p and the cellular-based V2X (C-V2X). The former relies on direct communication between vehicles using the wireless access in vehicular environmental (WAVE) protocol. On the other hand, C-V2X, which is based on cellular telecommunications, achieves V2X via direct device peering and/or through the cellular operators [1], [2]. WAVE under IEEE 802.11p is to operate within the Wi-Fi band of 5.77–5.925 GHz, whereas C-V2X is to operate vis-à-vis legacy supported third Generation Partnership Project cellular bands. These include but are not limited to 3G, 4G long-term evolution (LTE), and the tentative IMT2020 5G bands between 600 and 3500 MHz. Due to the variety of operating bands and applications of V2X systems, be it IEEE 802.11p or C-V2X, a feasible measurement facility for testing the wireless performance of a vehicle within the range of 600 MHz to 6 GHz is needed. Fig. 1 illustrates a prototype of an inexpensive, simple to construct and rapid V2X communication measurement system. As can be seen, three identical antennas are mounted on a gantry (or wall) to test the signal reception at different grazing incidences of electromagnetic waves. With this, an existing vehicle testing facility can be utilized for V2X testing in the most economical manner. The mounted antennas should be Manuscript received February 27, 2018; revised November 27, 2018; accepted December 9, 2018. Date of publication January 21, 2019; date of current version March 5, 2019. This work was supported by the Seoul National University of Science and Technology through the Advanced Research Project. (Corresponding author: Jae-Young Chung.) The authors are with the Electrical and Information Department, Seoul National University of Science and Technology, Seoul 01811, South Korea (e-mail: jychung@seoultech.ac.kr). Color versions of one or more of the figures in this communication are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TAP.2019.2893201 compact, lightweight, and inexpensive and, above all, should have ultrawideband (UWB) radiation characteristics and dual-polarization capability. The use of quad-ridged horn antennas in [3]–[5] has been the traditional method of implementing dual-polarized wideband antennas. Although these offer excellent performance across the bandwidth, they tend to be bulky, heavy, and expensive to develop. Alternatively, dual-polarized Vivaldi antennas implemented with two orthogonally crossed printed circuit boards (PCB) [6]–[8] offer excellent radiation performances in a more economical way. However, a significant drawback of Vivaldi antenna is the poor impedance matching condition, low gains at lower frequencies and a midfrequency gain drop. The latter is present in the cross-shaped antennas only. As reported in [9] that for longer wavelengths (<2 GHz), the guided waves along the tapered edges reflect at the wider ends and form standing-waves; making low-frequency impedance matching difficult. This also results in the radiation pattern deformation and the antenna gain reduction. To correct this challenge, Han et al. [7] introduced corrugated ripples on the outer edges of the radiating arms on two cross-shaped Vivaldi antennas. It achieved a 140% fractional bandwidth (fBW) from 1.4 to 8.0 GHz. Also, Sonkki et al. [6] developed a pair of cross-shaped Vivaldi antennas with a 166% fBW at −10 dB S11 from 0.683 GHz up to 7.0 GHz. Using a traditional tapered balun structure, it achieved the wideband results mainly due to good optimization and a large antenna size of 220 mm × 240 mm. This communication was set out to overcome the above-mentioned challenge of the Vivaldi antenna by introducing a two-stage quarterwave balun at the feed, thus improving the low-frequency matching condition without increasing the size arbitrarily. The macroscopic air gaps between the cross-shaped Vivaldi antennas which also results in gain drops at the mid-frequency was investigated and corrected with silicone adhesive. The proposed design was fabricated, measured, and implemented in a V2X test system to ascertain the results. An fBW of 172.95% (0.56–7.72 GHz) is achieved. The realized gain of up 0018-926X © 2019 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information. Authorized licensed use limited to: Universidad de los Andes. Downloaded on May 23,2020 at 00:17:10 UTC from IEEE Xplore. Restrictions apply. 1958 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 67, NO. 3, MARCH 2019 Fig. 2. Two-stage quarter-wave balun. (a) Detailed geometry of balun. (b) Front/Back view of antenna. to 9.2 dB is also achieved at a compact size of 187.5 mm × 190 mm. These results are not only superior to the previously reported cross-shaped Vivaldi antennas [6]–[8] but also provide alternative approaches to improving the Vivaldi antenna’s shortcomings. With preliminary simulation and some measured results of the antenna discussed in a conference communication [10], this communication discusses details of the balun design and measured results. Moreover, using the proposed model as a test antenna, the radiation pattern measurement results of the roof-top shark antenna are obtained from the V2X testing facility (described in Fig. 1). These are presented in Section IV to demonstrate the usefulness of the proposed design. II. A NTENNA D ESIGN Fig. 3. (a) Real input impedance of balun by varying width w1 . (b) Imaginary input impedance of balun by varying width w1 . A. Design Procedure The Vivaldi antenna or tapered slot antenna (TSA) has a hypothetically unlimited instantaneous frequency bandwidth with stable high gains and linear polarization [9], [11]. Therefore, it is frequently used in wideband communication systems and testing facilities. Due to its planar structure, the antenna can be easily fabricated using the PCB technology with various modifications [12]. The geometry and dimensions of the antenna are shown in the inset of Fig. 1 [10]. The linear dual-polarization is achieved by orthogonally integrating the two identical Antennas A and B; which were fabricated on 0.8 mm thick FR-4 substrates. Two slits incisions were made from either ends of each antenna toward the feeding point to allow one antenna to be slid into the other. The exponentially tapered profiles (shaded region in Fig. 1) offer smooth impedance transition from guided waves (on the radiators) to radiated waves (in free space). The width of the tapered end defines the lowest cutoff frequency of the antenna at the guided wavelength. This width in Fig. 1 is 187.5 mm, corresponding to the guided wavelength at 750 MHz when FR-4 substrate with the dielectric constant (εr ) of 4.4 is used. Thus, instead of the intended 600 MHz opening gap, 750 MHz was used. This reduction was made possible in part by rigorous optimization of the radiator taper and the implementation of the quarter-wave balun feedline. The high-frequency cutoff region is also determined by the opposite end of the tapered structure, where the antenna is fed. Vivaldi antenna baluns are, therefore, designed with the mid-frequency wavelength; to ensure a balanced matching condition across the bandwidth of operation. With these methods, impedance performance at the lower frequency regions below 2 GHz was greatly improved, hence, the size reduction. B. Two-Stage Quarter-Wave Balun Feeding The Vivaldi guides the traveling wave along its balanced structure and thus requires a balanced-to-unbalanced (balun) transformer when fed by an unbalanced coaxial cable. Fig. 2 shows the implemented two-stage quarter-wave balun structure at the Vivaldi feed. The rightangled balun feedline at the backside of the substrate, couples with the radiators at the narrow slot of width w1 just above the circular cavity. The purpose of employing the quarter-wave balun instead of a widely used tapered balun [6], [12] is to improve the bandwidth. The impedance transition stages of the quarter-wave balun are from the 50 input (length l1 ) through to the open circuit (OC) radial stub. The slot impedance of 138 is calculated at the center frequency ( f C ) of 3.3 GHz for the slot width w1 . This width is kept very narrow to achieve better impedance matching conditions at higher frequencies (>3 GHz). However, at narrower w1 , higher capacitances are introduced into the balun, which can be suppressed by extending l4 and/or r1 (i.e., increasing the inductance of the resonator). Nevertheless, this is nearly impossible due to the orthogonal cross-integration limitations of the Vivaldi; as it can cause galvanic contact between the two antennas. Another challenge is that the impedance of the feedline at the coupling point must be high enough to properly couple with the high slot impedance. This requires a very thin microstrip feedline width at this point; which is also limited by fabrication precision and accuracy. The slot width is, therefore, widened to reduce the high capacitances and high slot impedance. The increment of w1 results in mid-frequency gain drops which have been discussed in Section III of this communication. Due to these constraints, the size of cross-shaped Vivaldi antennas is often enlarged beyond the cutoff frequency to maintain the wideband impedance matching and compensate for gain drops. These limitations only leave room for rigorous feed design and optimization. The two-stage quarter-wave balun was designed using the Klopfenstein taper approach in [13] and [14]. A full-wave EM simulation software, Ansys HFSS [15], was used to optimize the balun geometry. The optimization was performed by observing the input impedances and S11 while varying w1 , r2 , and l2 among other parameters. Fig. 3(a) and (b) shows the real and imaginary input impedances when w1 is varied. In this simulation, the OC radial stub is replaced by a lumped port to optimize the performance of the slot width alone. The lower frequency band (<1 GHz) hardly changes by varying Authorized licensed use limited to: Universidad de los Andes. Downloaded on May 23,2020 at 00:17:10 UTC from IEEE Xplore. Restrictions apply. IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION 1959 TABLE I O PTIMIZED BALUN PARAMETERS Fig. 4. Input impedance of the balun by varying r2 . Fig. 5. S11 by varying length l2 ≈ l3 . w1 for both real and imaginary components. At higher frequencies, however, the real input impedance, which properly characterizes the slot, is sensitive to w1 . A large fluctuation is observed in both the real and imaginary curves at around 6.0 GHz. This is due to the unmatched OC port used, which defines the high-frequency cutoff region. This cutoff is usually extended with the radial stub technique. Although w1 = 1 mm in Fig. 3 is better for 50 matching; it might cause galvanic contact with the second antenna due to the 0.8 mm substrate thickness. It was, therefore, optimized to 1.65 mm, which is more twice the substrate thickness. Fig. 4 also shows the absolute input impedance by varying r2 , the radius of the radial stub. The real/imaginary components are not shown since the optimized stub effectively removes the reactance components in the feedline; hence, making the real components similar to the absolute value. The increase in impedance is clearly observed as r2 increases at the lower frequency of 500 MHz. The radial stub at certain lengths reaches antiresonance as its dominant inductive property reverts to capacitive. This change makes the stub an OC and is effective for canceling the remaining currents in the feedline. At r2 = 6.7 mm, the frequency resonance is seen to be closer to the 50 line; indicating good matching. Also, the cutoff region is extended beyond 6 GHz by the stub; as the fluctuations observed around 6 GHz in Fig. 3 are reduced. Fig. 5 shows the final S11 optimization by varying l2 and l3 simultaneously. As can be seen, it greatly affects the impedance matching characteristics across the bandwidth and dominantly around the mid-frequency region. l2 was, therefore, set at 11.68 mm ≈ 12 mm, which shows the best result. This confirms the impedance transition with two quarter-wavelength stages for λg /4 = 11.68 mm at f C = 3.3 GHz. Their (l2 and l3 ) equivalent widths were set to 0.75 and 0.3 mm corresponding to ∼73 and ∼106 . The optimized values are summarized in Table I. III. A NTENNA FABRICATION AND M EASUREMENT A. Cross-Integration and Mid-Frequency Gain Drop Fig. 6 shows the fabricated and assembled antennas. The frontside of Antenna A with the longitudinal slit (S1) is shown in Fig. 6(a). The Fig. 6. Fabricated Vivaldi antenna pair. (a) Front-side of Antenna A. (b) Back-side of Antenna B. (c) Assembled antenna with silicone adhesive (left inset) and soldering (right inset). inset shows the circular cavity and the thin microstrip line connecting the two radiator halves. The backside of the second Antenna B is shown in Fig. 6(b) with the quarter-wave balun and second slit (S2). The two antennas are slid into each other with the base of Antenna 2 soldered as shown in the inset of Fig. 6(c) (right). The soldering connects the separating halves of Antenna B. To ensure rigidity and measurement stability, the two antennas are encased in a transparent holder. The case did not, however, affect the electrical performance of the antenna. During fabrication and measurement process, the cross-shaped Vivaldi antenna had a structural limitation which reduced the midfrequency gain values. As discussed in Section II and from Fig. 2, the slit (S1) in Antenna A was optimized to 1.65 mm to ensure good impedance matching with the balun, prevent galvanic contact and minimize high capacitance in the balun. However, with a substrate thickness of 0.8 mm, an air gap of 0.85 mm is created when both antennas are cross-shaped. The air gap tends to introduce more resonant interferences between the antennas at the middle frequency which results in gain losses. References [6] and [7] among other cross-shaped Vivaldi references also reported gain drops between 1.5 and 2.5 dB at their midfrequencies; making the midfrequency gain drop an inherent drawback in the cross-shaped Vivaldi antenna. Most researchers use larger radiator Authorized licensed use limited to: Universidad de los Andes. Downloaded on May 23,2020 at 00:17:10 UTC from IEEE Xplore. Restrictions apply. 1960 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 67, NO. 3, MARCH 2019 Fig. 8. Simulated and measured S11 and port isolation. Fig. 9. Simulated and measured realized gain. Fig. 7. Comparison of antenna with (w.) and without (w/t.) silicone adhesive and styrofoam sealant. (a) Measured gain (port 1). (b) Measured S11. sizes to compensate for the gain drop. From Fig. 7(a), two resonant dips of up to 3 dB are observed at around 3.0 and 5.8 GHz for the antenna with air gaps. These dips are, however, nonexistent in the single antenna measurement and less pronounced in the simulation results. The gain drop is not suitable for an antenna in a UWB test system. The authors, therefore, experimented with various novel approaches to mitigate the gain drop problem. These included sealing the air gaps with dissolved styrofoam or with silicone adhesive, having a low dielectric permittivity. Alternatively, the crossed Vivaldi structure can be constructed using 3-D printing. The latter is, however, complex and not practical for this design. The left-inset photograph in Fig. 6(c) shows the antenna with silicone adhesive (silicone 1035, εr = 2.8), used to seal the air gaps. Minimal amounts were carefully applied at the intersecting regions without touching the radiators. The same sealing technique was applied with the styrofoam paste made by dissolving a piece of styrofoam in pure acetone. After about a 12 h waiting time, the paste hardens and is secure. Fig. 7 shows the measured results of the Vivaldi antenna with and without the silicone and styrofoam paste. Silicone 1055, which is widely used for bonding circuit components, offered better performance with no air gaps after sealing. styrofoam, on the other hand, still had some trapped air which resulted in a poorer performance compared with the silicone. Gain improvement of about 2.1 dBi was recorded for the styrofoam paste model in Fig. 7(a). The best performance was observed with the silicone sealant. It achieved gain improvement of about 2.8 dBi. Similar results are observed in the S11 of Fig. 7(b) with the styrofoam and silicone adhesives performing better than the air-gap model. As noted, the silicone sealant with a lower dielectric constant did not Fig. 10. Simulated and measured XPD. alter the antenna’s intrinsic parameters as the results (both S11 and far-field patterns) were similar to the simulated values. B. Measured Result Discussion Fig. 8 shows the simulated and measured S11, S22, and S21. For the measured results, an fBW of 172.95% is recorded at −10 dB S11 (560 MHz to 7.72 GHz) for Antenna A (Port 1). Also, the port isolation (S21) of better than −28 dB is observed. A similar S11 is obtained from Antenna B (Port 2), where a 173.5% fBW at −10 dB S22 (530 MHz to 7.48 GHz) is obtained. Also, at −8 dB, a 173% and 175.3% fBW for S11 and S22 are observed, respectively. The measured results agreed well with the simulation results. The fabricated antenna was brought into an anechoic chamber for the measurements of radiation characteristics. The gradual change in gain levels across the bandwidth is a common phenomenon with traveling wave structures, that is, their gains are proportional to the antenna size and the rate at which the energy dissipates as it travels along the tapered structure. Fig. 9 shows the realized gains ranging from 1.2 to 9.2 dBi across the bandwidth. Although the structural minimization also affected the gain performance, the quarter-wave balun enabled the minimal gain increment of 1.2 dBi at the lowest frequency. Fig. 10 shows the cross-polarization discrimination (XPD) which is better than 17.2 dB for both ports, when measured. Although there is a change in the XPD of the simulated and measured results, 17.2 dB XPD is acceptable for a test antenna. Normalized radiation patterns in decibels are shown in Fig. 11. Similar radiation patterns between the ports 1 and 2 antennas are observed. The WAVE Authorized licensed use limited to: Universidad de los Andes. Downloaded on May 23,2020 at 00:17:10 UTC from IEEE Xplore. Restrictions apply. IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION 1961 Fig. 11. Normalized radiation patterns of measured co-polarization (Cpol.) and cross polarization (Xpol.). (a) 0.7 GHz. (b) 1.0 GHz. (c) 3.0 GHz. (d) 5.8 GHz. TABLE II C OMPARISON OF P ROPOSED M ODEL W ITH S IMILAR D ESIGNS frequency of 5.8 GHz is also shown to have very good performance. An average 3 dB beamwidth ranging between 77.2° and 45.3° for port 1 and 88.7°–37.9° for port 2 is also observed across the bandwidth. Table II summarizes the performance of the proposed antenna and compares it with other reports on the cross-shaped Vivaldi antenna design. Evidently, the proposed design has better results compared with the existing structures with respect to structural compactness, bandwidth enhancement and gain stability. IV. V2X S YSTEM With the required antenna performance for the V2X system achieved, this section discusses the measurement results in the V2X test facility. A. System Setup At an indoor C-V2X test facility, the antennas are configured as seen in Figs. 1 and 12. The latter figure shows a partial photograph of the measurement facility with the three-antenna array. Antennas (Ant.) 1, 2, and 3 are mounted on a gantry at 50°, 60°, and 70° grazing incident angles. These are represented as channels 1, 2, and 3 in the radiation patterns of Fig. 13. Each antenna is encased in a special radome to protect it and reduce its sensitivity. This is to mimic unfavorable outdoor conditions at the interface of the transceiver antennas for worst case scenarios. The shark antenna used for the test is also shown in the inset of Fig. 12. This is a proprietary MIMO transceiver shark antenna mounted on a vehicular platform and operating within the standard cellular frequency band. Due to system limitations at the measurement facility, the IEEE 802.11p standard which includes the WAVE frequency band was not tested. The antenna, however, is designed to operate in these bands as well; as seen from the measured radiation pattern in Fig. 11(c) and (d). Therefore, the following bands were under test for C-V2X in the facility. GSM at 850–1900 MHz with related PCS and DCS systems, 7002100 MHz for UMTS-based communication Fig. 12. V2X measurement chamber with three mounted Vivaldi antennas. Inset: rooftop mounted shark antenna. variants and 600–3500 MHz for 4G-LTE systems. At these cellular enabled frequency bands, the mounted TSAs served as an ideal transceiver in a stationary scenario using a proprietary cellular network configuration. Channel 1 represented vehicle-to-infrastructure and vehicle-to-network systems as these have ideally similar interface structures. The antenna is placed at a 50° inclined angle from the normal of the shark antenna. Vehicle-to-vehicle is channel 2° at 60 from the vehicle and channel 3° at 70 is vehicle-to-pedestrian scenario. B. Measurement Results Fig. 13 shows the received gains of the shark antenna at certain frequency bands with the Vivaldi antennas. Fig. 13(a) shows 915 MHz for the 900 MHz GSM band (15 MHz bandgap required). The results show a fairly good isolation between the horizontal and vertical patterns; representing the two linear polarizations of the Vivaldi antenna. The 900 MHz GSM band has receiver gain (Rx. G) of up to 5 dBi across all three channels. Similarly, Fig. 13(c) at 787 MHz LTE frequency-division multiplexed (FDD) has a relatively good polarization isolation with Rx. G of up to 3.3 dBi. The 2100 UMTS band, recorded at 2110 MHz in Fig. 13(b), has Rx. G of up to 7 dBi but with a much irregular receive pattern in channel 2 as the horizontal polarization is seen to be stronger than expected. The LTE timedivision multiplexed (TDD) patterns are also observed in Fig. 13(d) at 1930 MHz with similar characteristics as the 2100 UMTS. Authorized licensed use limited to: Universidad de los Andes. Downloaded on May 23,2020 at 00:17:10 UTC from IEEE Xplore. Restrictions apply. 1962 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 67, NO. 3, MARCH 2019 Fig. 13. Measured radiation patterns by the Vivaldi mounted antennas. (a) GSM 900 MHz band. (b) UMTS/WCDMA band 2100 MHz. (c) FDD LTE 700 MHz. (d) TDD LTE 2100 MHz. In all, the received gains across the three scenarios proved stable and efficient for the measurement. The overall effect of the quarterwave balun is seen as being superior in impedance matching as compared with the traditional approach; as it offered the opportunity to measure antennas at these frequencies in a cost-effective manner. V. C ONCLUSION A compact C-V2X Vivaldi test antenna has been proposed with up to 172.95% fBW from 560 MHz to 7.72 GHz. A two-stage balun design has been put forward in the Vivaldi antenna design to improve a known drawback of the antenna: low-frequency input impedance matching. A simple silicone adhesive sealing technique has also been investigated to correct the mid-frequency gain drop in cross-shaped Vivaldi antennas. This structure also achieved good gains in the lowfrequency region of the antenna. Although more can be done in later studies to improve the low-frequency gains, the antenna has sufficient gain stability for measuring UWB V2X communication standards in a controlled environment. Measurement results from the antenna in the V2X test facility have also been presented with stable results. R EFERENCES [1] A. Papathanassiou and A. Khoryaev, “Cellular V2X as the essential enabler of superior global connected transportation services,” IEEE 5G Tech Focus, vol. 1, no. 2, p. 2017, Jun. 2017. [2] K. Wevers and M. Lu, “V2X communication for ITS-From IEEE 802.11p towards 5G,” IEEE 5G Tech Focus, vol. 1, no. 2, Jun. 2017. [Online]. Available: https://futurenetworks.ieee.org/tech-focus/march2017/v2x-communication-for-its [3] S. Soroka, “A physically compact quad ridge horn design,” in Proc. IEEE Antennas Propag. Soc. Int. Symp., Philadelphia, PA, USA, Jun. 1986, pp. 903–906. [4] G. M. Coutts, “Wideband diagonal quadruple-ridge orthomode transducer for circular polarization detection,” IEEE Trans. Antennas Propag., vol. 59, no. 6, pp. 1902–1909, Jun. 2011. [5] A. Akgiray, S. Weinreb, W. A. Imbriale, and C. Beaudoin, “Circular quadruple-ridged flared horn achieving near-constant beamwidth over multioctave bandwidth: Design and measurements,” IEEE Trans. Antennas Propag., vol. 61, no. 3, pp. 1099–1108, Mar. 2013. [6] M. Sonkki, D. Sánchez-Escuderos, V. Hovinen, E. T. Salonen, and M. Ferrando-Bataller, “Wideband dual-polarized cross-shaped Vivaldi antenna,” IEEE Trans. Antennas Propag., vol. 63, no. 6, pp. 2813–2819, Jun. 2015. [7] X. Han, L. Juan, C. Changjuan, and Y. Lin, “UWB dual-polarized Vivaldi antenna with high gain,” in Proc. Int. Conf. Microw. Millim. Wave Technol. (ICMMT), May 2012, pp. 1–4. [8] J. Zhang, E. C. Fear, and R. H. Johnston, “Cross-Vivaldi antenna for breast tumor detection,” Microw. Opt. Technol. Lett., vol. 51, no. 2, pp. 275–280, Feb. 2009. [9] P. J. Gibson, “The Vivaldi aerial,” in Proc. 9th Eur. Microw. Conf., Sep. 1979, pp. 101–105. [10] P. A. Dzagbletey, J. Y. Shim, J. Y. Jeong, and J.-Y. Chung, “Dualpolarized Vivaldi antenna with quarter-wave balun feeding,” in Proc. Asia–Pacific Int. Symp. Electromagn. Compat. (APEMC), Seoul, South Korea, Jun. 2017, pp. 108–110. [11] J. Bang, J. Lee, and J. Choi, “Design of a wideband antipodal Vivaldi antenna with an asymmetric parasitic patch,” J. Electromagn. Eng. Sci., vol. 18, no. 1, pp. 29–34, Jan. 2018. [12] S. Hwang, B. Lee, D. H. Kim, and J. Y. Park, “Design of S-band phased array antenna with high isolation using broadside coupled split ring resonator,” J. Electromagn. Eng. Sci., vol. 18, no. 2, pp. 108–116, Apr. 2018. [13] D. M. Pozer, “Impedance matching and tuning,” in Microwave Engineering. Hoboken, NJ, USA: Wiley, 2012, pp. 264–267. [14] B. G. Evans, “Chebyshev quarter-wave stepped balun transformer,” Electron. Lett., vol. 9, no. 22, pp. 514–516, Nov. 1973. [15] Electronics Desktop, High Frequency Structure Simulator (HFSS), Release 18.0, ANSYS, Canonsburg, PA, USA. [Online]. Available:https://www.ansys.com/academic/terms-and-conditions Authorized licensed use limited to: Universidad de los Andes. Downloaded on May 23,2020 at 00:17:10 UTC from IEEE Xplore. Restrictions apply.
