EE214 Advanced Analog IC Design-course Murmann

EE214 Advanced Analog Integrated Circuit Design - Winter 2011 Boris Murmann Stanford University murmann@stanford.edu Table of Contents Chapter 1 Chapter 2 Chapter 3 Chapter 4 Chapter 5 Chapter 6 Chapter 7 Chapter 8 Chapter 9 Chapter 10 Introduction BJT Devices MOS Devices and Gm/ID-based Design Review of Elementary Circuit Configurations Two-Port Feedback Circuit Analysis Frequency Response of Feedback Circuits Wideband Amplifiers Noise Analysis Distortion Analysis Opamps and Output Stages Chapter 1 Introduction B. Murmann Stanford University Analog Circuit Sequence Fundamentals for upperlevel undergraduates and graduate entry-level y g students Analysis and design techniques for highperformance circuits in p advanced Technologies Design of mixedsignal/RF building bl k blocks EE314 — RF Integrated Circuit Design EE114 — Fundamentals of Analog Integrated Circuit Design EE214 — Advanced Analog Integrated Circuit Design EE315A — VLSI Signal Conditioning Circuits EE315B — VLSI Data Conversion Circuits B. Murmann EE214 Winter 2010-11 – Chapter 1 2 The Evolution of a Circuit Designer… EE101A,B B. Murmann EE114 EE214 EE314 EE315A,B EE214 Winter 2010-11 – Chapter 1 3 Analog Design EE 114 Bandwidth Power Dissipation EE 214 Electronic Noise B. Murmann EE214 Winter 2010-11 – Chapter 1 Distortion 4 Significance of Electronic Noise (1) Signal-to-Noise g Ratio SNR = B. Murmann Psignal Pnoise EE214 Winter 2010-11 – Chapter 1 ∝ 2 Vsignal 2 Vnoise 5 Significance of Electronic Noise (2) Example: Noisy image http://www.soe.ucsc.edu/~htakeda/kernelreg/kernelreg.htm p g g B. Murmann EE214 Winter 2010-11 – Chapter 1 6 Significance of Electronic Noise (3) Th The "fidelity" "fid lit " off electronic l t i systems t is i often ft determined d t i d by b their th i SNR – Examples • Audio systems • Imagers, Imagers cameras • Wireless and wireline transceivers Electronic noise directly trades with power dissipation and speed – In I mostt circuits, i it low l noise i di dictates t t llarge capacitors it ((and/or d/ smallll R R, large gm), which means high power dissipation Noise has become increasingly important in modern technologies with reduced supply voltages – SNR ~ Vsignal2/Vnoise2 ~ (αVDD)2/Vnoise2 Designing a low-power, high-SNR circuit requires good understanding of electronic l t i noise i B. Murmann EE214 Winter 2010-11 – Chapter 1 7 Distortion vo vi Small-signal approximation All electronic circuits exhibit some level of nonlinear behavior – The resulting waveform distortion is not captured in linearized smallsmall signal models The distortion analysis tools covered in EE214 will allow us to quantify the impact of nonlinearities on sinusoidal waveforms B. Murmann EE214 Winter 2010-11 – Chapter 1 8 Significance of Distortion For a single tone input, the nonlinear terms in a circuit’s transfer function primarily result in signal harmonics For a two-tone input, the nonlinear terms in a circuit’s transfer function result lt in i so-called ll d “i “intermodulation t d l ti products” d t ” Example: Two interferer tones create an intermodulation product that corrupts the signal in a desired (radio-) channel B. Murmann EE214 Winter 2010-11 – Chapter 1 9 Noise and Distortion Analysis in EE214 Main objective – Acquire the basic tools and intuition needed to analyze noise and distortion in electronic circuits – Look at a few specific circuit examples to “get a feel” for situations where noise and/or distortion may matter Leave application-specific examples for later – EE314: Noise and distortion in LNAs,, mixers and power p amplifiers p – EE315A: Noise and distortion in filters and sensor interfaces – EE315B: Noise and distortion in samplers, A/D & D/A converters B. Murmann EE214 Winter 2010-11 – Chapter 1 10 Technology MOSFET Bipolar Junction Transistor (BJT) EE114 EE214 B. Murmann EE214 Winter 2010-11 – Chapter 1 11 Bipolar vs. CMOS (1) Advantages of bipolar transistors – Lower parametric variance – Higher Hi h supply l voltages l – Higher intrinsic gain (gmro) – Higher fT for a given feature size/lithography Disadvantages of bipolar transistors – Lower integration density, larger features – Higher cost B. Murmann EE214 Winter 2010-11 – Chapter 1 12 Bipolar vs. CMOS (2) A.J. Joseph, et al., "Status and Direction of Communication Technologies - SiGe BiCMOS and RFCMOS," Proceedings of the IEEE, vol.93, no.9, pp.1539-1558, September 2005. • CMOS tends to require finer lithography to achieve same speed as BiCMOS p process with advanced BJT B. Murmann EE214 Winter 2010-11 – Chapter 1 13 Example that Leverages High-Speed BJTs: 40Gb/s Integrated Optical Transponder B. Murmann EE214 Winter 2010-11 – Chapter 1 14 Die Photo of 40Gb/s CDR Circuit * • 120-GHz fT / 100GHz fmax 0.18μm SiGe BiCMOS • 144 pins • 3.5mm x 4.2mm • +1.8V and –5.2V supplies • 7.5W power dissipation *A A. O Ong, ett al., l ISSCC 2003 B. Murmann EE214 Winter 2010-11 – Chapter 1 15 Radar Sensor R Ragonese ett al., l ISSCC 2009 B. Murmann EE214 Winter 2010-11 – Chapter 1 16 Example that Leverages Densely Integrated CMOS: RF Transceiver System-on-a-Chip (SoC) In modern CMOS technology, millions of l i gates logic t can b be integrated on a chip – Together with moderate- to high moderate performance analog blocks Mehta et al., "A 1.9GHz Single-Chip CMOS PHS Cellphone," ISSCC 2006. B. Murmann EE214 Winter 2010-11 – Chapter 1 17 45nm CMOS (Intel) Steve Cowden THE OREGONIAN July 2007 B. Murmann EE214 Winter 2010-11 – Chapter 1 18 Research in Device Technology Vdd A Transport-enhanced FET Molecular devices B Out Strained Si, Ge, SiGe, III-V A B buried oxide Gnd isolation Silicon Substrate Self-assembled device fabrication Nanodevice array Spintronics top-gate channel Nanotube 3D, heterogeneous i t integration ti channel Quantum cascade back-gate isolation buried oxide Multi-Gate / FinFET Nanowire Gate Source Drain [Prof. P. Wong] Embedded memory Ferromagnetic domain wall Time B. Murmann EE214 Winter 2010-11 – Chapter 1 19 Thoughts on Device Technology In the future, innovative circuit designers must embrace “whichever” technology is most suitable (in terms of performance, cost, reliability, etc. f their for th i specific ifi problem bl – Regardless of the respective I-V law and associated nonidealities In EE214, we will use bipolar and MOS technology to illustrate the similarities and differences between two advanced technologies The device parameters and simulation models of the “EE214 EE214 technology” correspond to a modern 0.18-μm SiGe BiCMOS technology – See e.g. S. Wada, et al., “A manufacturable 0.18-um SiGe BiCMOS technology for 40-Gb/s optical communication LSIs,” in Proc. 2002 Bipolar/BiCMOS Circuits and Technology Meeting, pp. 84–87. B. Murmann EE214 Winter 2010-11 – Chapter 1 20 Analysis of Feedback Circuits F Feedback db k circuits i it can b be studied t di d iin severall ways – Return ratio analysis (EE114) – Two-port analysis (EE214) Both methods have their own merits and demerits, and a good circuit designer should understand both approaches Two-port analysis nicely captures a number of practical scenarios in which the forward amplifier (“a”) and feedback network (“f”) can be intuitively identified and separated (while maintaining loading effects) – Shunt-shunt, shunt-series, series-shunt, series-series configurations Example: Shunt series feedback circuit Shunt-series B. Murmann EE214 Winter 2010-11 – Chapter 1 21 Root Locus Techniques Provides intuitive guidance on “where the poles move” when the loop gain is varied Valuable for stability analysis and frequency compensation B. Murmann EE214 Winter 2010-11 – Chapter 1 22 Course Outline BJT & short channel MOS device models Review of elementaryy amplifier p stages g ((BJT focus)) Two-port feedback circuit analysis Root locus Wideband amplifiers Noise analysis Distortion analysis OpAmps and output stages B. Murmann EE214 Winter 2010-11 – Chapter 1 23 Summary of Learning Goals Understand device behavior and models for transistors available in advanced integrated circuit technologies – SiGe BJT, short channel MOS Acquire the basic intuition and models for – Distortion analysis – Noise analysis – Two-port feedback circuit analysis – Root locus techniques and their application to broadband amplifiers Solidify the above topics in a hands-on project involving the design and optimization of a broadband amplifier circuit B. Murmann EE214 Winter 2010-11 – Chapter 1 24 Staff and Website Instructors – Boris Murmann, Drew Hall Teaching assistants – Kamal Aggarwal, Pedram Lajevardi Administrative support – Ann Guerra, CIS 207 Lectures are televised – But please come to class to keep the discussion interactive! Web page: http://ccnet.stanford.edu/ee214 – Check regularly, especially bulletin board – Register for online access to grades and solutions B. Murmann EE214 Winter 2010-11 – Chapter 1 25 Text and Prerequisites EE214 C Course reader d – Hardcopies available at Stanford Bookstore (~1/3) Required textbook – Gray, Hurst, Lewis and Meyer, Analysis and Design of Analog Integrated Circuits, 5th ed., Wiley Reference text – B. Razavi, Design of Integrated Circuits for Optical Communications, McGraw-Hill, 2002 Course p prerequisite: q EE114 or equivalent q – Basic device physics and models – Frequency response, dominant pole approximation, ZVTC – Biasing, g, small-signal g models – Common source, common gate, and common drain stages – Port impedance calculations – Feedback basics B. Murmann EE214 Winter 2010-11 – Chapter 1 26 Assignments Homework (20%) – Handed out on Wed, due following Wed in class – Lowest L t HW score will ill b be d dropped d – Policy for off-campus students • Fax or email to SCPD before deadline stated on handout Midterm Midt E Exam (30%) Design Project (20%) – Design of an amplifier using HSpice (no layout) – Work W k in i tteams off two t – OK to discuss your work with other teams, but no file exchange! Final Exam (30%) B. Murmann EE214 Winter 2010-11 – Chapter 1 27 Honor Code Please remember you are bound by the honor code – We will trust you not to cheat – We will try not to tempt you But if you are found cheating it is very serious – There is a formal hearing g – You can be thrown out of Stanford Save yourself a huge hassle and be honest For more info – http://www.stanford.edu/dept/vpsa/judicialaffairs/guiding/pdf/honorcode.pdf B. Murmann EE214 Winter 2010-11 – Chapter 1 28 Chapter 2 Bipolar p Junction Transistors B. Murmann Stanford University Reading Material: Sections 1.1, 1.2, 1.3, 1.4, 2.5, 2.6, 2.7, 2.11, 2.12 History Bardeen, Brattain, and Shockley, 1947 W. Brinkman, D. Haggan, and W. Troutman, “A history of the invention of the transistor and where it will lead us,” IEEE J. Solid-State Circuits, vol. 32, no. 12, pp. 1858-1865, Dec. 1997. W. Shockley, W Shockley M. M Sparks, Sparks and G G. K K. Teal Teal, “P P-N N junction transistors,” Phys. Rev. 83, pp. 151–162, Jul. 1951. B. Murmann EE214 Winter 2010-11 – Chapter 2 2 Conceptual View of an NPN Bipolar Transistor (Active Mode) IC ∝ e IB << IC VBE B. Murmann n- C p B n+ E Device acts as a voltage controlled current source – VBE controls IC VBE kT / q The base-emitter junction is forward biased and the basebase collector junction is reverse biased VCE The device is built such that – The base region is very thin – The emitter doping is much higher than the base doping – The collector doping is much lower than the base doping EE214 Winter 2010-11 – Chapter 2 3 Outline of Discussion In order to understand the operation principle of a BJT, we will look at – The properties of a forward biased pn+ junction – The properties of a reverse biased pn- junction – And the idea of combining the two junctions such that they are joined by a very thin (p-type) base region The treatment in the following slides is meant to be short and qualitative – See any solid-state physics text for a more rigorous treatment (involving band diagrams, etc.) B. Murmann EE214 Winter 2010-11 – Chapter 2 4 pn+ Junction in Equilibrium (No Bias Applied) nn0 ≅ ND (Donor concentrat ion) pp0 ≅ NA (Acceptor concentrat ion) nn pn np pp np 0 = ni2 n2 ≅ i p p 0 NA pn0 = ni2 n2 ≅ i nn0 ND Concentration of electrons on n side (majority carriers) Concentration of holes on n side (minority carriers) Concentration of electrons on p side ((minority y carriers)) Concentration of holes on p side (majority carriers) The subscript “0” in the carrier concentrations denotes equilibrium (no bias applied) B. Murmann EE214 Winter 2010-11 – Chapter 2 5 Built-in Potential The built in potential sets up an electric field that opposes the diffusion of mobile holes and electrons across the junction (Drift ) qμppE = qDp dp dx (Diffusion) ⎛n ⎞ ⎛p ⎞ ⎛N N ⎞ ⇒ ψ 0 = VT ln⎜⎜ p0 ⎟⎟ = VT ln⎜ n0 ⎟ ≅ VT ln⎜⎜ A 2 D ⎟⎟ ⎟ ⎜ ⎝ ni ⎠ ⎝ pn0 ⎠ ⎝ np0 ⎠ B. Murmann EE214 Winter 2010-11 – Chapter 2 VT = kT q 6 pn+ Junction with Forward Bias (1) Depletion region narrows, diffusion processes are no longer balanced by electrostatic force At the edge of the depletion region (x=0), the concentration of minority carriers [np(0)] can be computed as follows ψ 0 − VBE B. Murmann ⎛ n ⎞ ⎛ N ⎞ = VT ln ⎜ n ⎟ ≅ VT ln ⎜ D ⎟ ⎜ n (0) ⎟ ⎜ n (0) ⎟ ⎝ p ⎠ ⎝ p ⎠ ∴np (0) = ND e ψ0 VT ⋅e VBE VT = np0 e VBE VT V n2 BE ≅ i e VT NA EE214 Winter 2010-11 – Chapter 2 7 pn+ Junction with Forward Bias (2) Th The result lt on the th previous i slide lid shows h th thatt fforward d bi biasing i iincreases the concentration of electrons at the “right” edge of the depletion region by a factor of exp(VBE/VT) The Th same h holds ld ffor h holes l att th the “l “left” ft” edge d off th the d depletion l ti region i VBE pn (0) = pn0 ⋅ e VT ≅ VBE ni2 ⋅ e VT ND Since ND >> NA, it follows that pn(0) << np(0), i.e. the concentration of minority carriers is much larger at the lightly doped edge Since there must be charge neutrality in the regions outside the depletion region, the concentration of the majority carriers at the edge of the depletion region must also increase − However, this increase is negligible when np(0) << pp ≅ NA (or pn(0) << nn ≅ ND) − These conditions are called “low-level injection” B. Murmann EE214 Winter 2010-11 – Chapter 2 8 What Happens with the Injected Minority Carriers? Th The carriers i would ld “like” “lik ” tto diff diffuse ffurther th iinto t th the neutral t l regions, i b butt quickly fall victim to recombination The number of minority carriers decays exponentially, and drops to 1/e off the th att the th so-called ll d diff diffusion i llength th (Lp or Ln, on the th order d off microns) i ) In each region, there are now two types of currents − Diffusion of injected j minority y carriers due to non-zero dnp/dx ((or dp pn/dx)) − Majority carrier currents for recombination n+ Total Current Jn Jp B. Murmann p Jp Jn Hole current (recombination) Electron diffusion current ( dnp/dx) EE214 Winter 2010-11 – Chapter 2 9 Summary – Forward Biased pn+ junction Lots of electrons being injected into the p-region, not all that many holes get injected into the n+ region – The heavier n-side doping, the more pronounced this imbalance becomes The electrons injected in the p region cause a diffusion current that d decays iin th the x-direction di ti d due tto recombination bi ti The recombination necessitates a flow of holes to maintain charge neutrality; as the diffusion current decays, the hole current increases, yielding i ldi a constant t t currentt d density it along l th the d device i Near the edge of the depletion region, the electron diffusion current dominates over the hole current that supplies carriers for recombination – This is a very important aspect that we will come back to B. Murmann EE214 Winter 2010-11 – Chapter 2 10 Reverse Biased pn- Junction Reverse bias increases the width of the depletion region g and increases the electric field Depletion region extends mostly into n- side Any electron that would “somehow” make it into the depletion region will be swept through, into the nregion – Due to electric field E eText, p.2 B. Murmann EE214 Winter 2010-11 – Chapter 2 11 Bipolar Junction Transistor – Main Idea n+ Total Current Jn Jp p Jp Hole current (recombination) Jn Electron diffusion current ( dnp/dx) “cut here” Make the p-region of the pn+ junction very thin Attach an n- region g that will “collect” and sweep p across most of the electrons before there is a significant amount of recombination B. Murmann EE214 Winter 2010-11 – Chapter 2 12 Complete Picture n+ p n- Text, p. 9 Straight line because base is thin; negligible recombination ((“short short base” base electron profile) B. Murmann EE214 Winter 2010-11 – Chapter 2 13 BJT Currents http://en.wikipedia.org/wiki/Bipolar p p g p _jjunction_transistor Primary current is due to electrons captured by the collector T Two (undesired) ( d i d) b base currentt components t – Hole injection into emitter (Æ 0 for infinite emitter doping) – Recombination in the base (Æ 0 for base width approaching zero) B. Murmann EE214 Winter 2010-11 – Chapter 2 14 First-Order Collector Current Expression Jn = qDn dnp (x) IC ≅ qADn dx ≅ −qDn np (0) WB np (0) Current density A is the cross-sectional area WB is the base width WB V n2 BE np (0) i e VT NA Result from slide 7 V qADnni2 VBET ∴ IC ≅ e WBNA ∴ IC ≅ IS e VBE VT B. Murmann IS = q nni2 qAD WBNA EE214 Winter 2010-11 – Chapter 2 15 Base Current IB = IB1+ IB2 where IB1 = Recombination in the base IB2 = Injection into the emitter IB1 follows f ll ffrom dividing di idi th the minority i it carrier i charge h iin th the b base (Qe) b by it its “lifetime” (τB) 1 VBE Qe 2 np (0)WBqA 1 WBqAni2 VT = = IB1= e τb τb 2 τbNA IB2 depends on the gradient of minority carriers (holes) in the emitter. For a long emitter (all minority carriers recombine) “long” ⎡ ⎛ 2 VBE − x dpn (x) d n L = −qADp ⎢ ⎜ i e VT e p IB2= −qADp ⎢ dx ⎜ N dx x =0 ⎢⎣ ⎝⎜ D VBE ⎞⎤ qADp ni2 V ⎟⎥ = e T ⎟⎟ ⎥ Lp ND ⎠ ⎥⎦ x =0 In modern narrow-base transistors IB2 >> IB1. B. Murmann EE214 Winter 2010-11 – Chapter 2 16 Terminal Currents and Definition of αF, βF Text, p. 9 IE = − (IC + IB ) βF IC IB αF IC β = F ( −IE ) 1 + βF (ideally infinite) (ideally one) Th The subscript b i t “F” iindicates di t th thatt th the d device i iis assumed d tto operate t iin th the forward active region (BE junction forward biased, BC reverse biased, as assumed so far) – More on other operating regions later later… B. Murmann EE214 Winter 2010-11 – Chapter 2 17 Basic Transistor Model Text, p. 13 Simplified model; very useful for bias point calculations (assuming e.g. VBE(on) = 0.8V) B. Murmann EE214 Winter 2010-11 – Chapter 2 18 Basewidth Modulation (1) Side note: BJT inherently has better (higher) ro than MOS since lower doping on n-side (collector) has most of the depletion region inside the collector Text, p. 14 VBE ∂IC ∂ ⎛ qADnni2 = e VT ⎜ ∂VCE ∂VCE ⎜⎝ WB (VCE ) ⋅ NA B. Murmann ⎞ I dWB ⎟=− C ⎟ W B dVCE ⎠ (See eq. (1.18) for dWB/dVCE term) EE214 Winter 2010-11 – Chapter 2 19 Early Voltage (VA) Text, p. 15 VA IC WB =− = const. (independent of IC ) ∂IC dWB dVCE ∂VCE IC ≅ ISe B. Murmann VBE VT ⎛ VCE ⎞ ⎜1+ ⎟ VA ⎠ ⎝ EE214 Winter 2010-11 – Chapter 2 20 Small-Signal Model V BE dIC I d gm = = ISe VT = C dVBE dVBE VT gπ = dIB 1 = rπ dVBE ⎛I ⎞ d⎜ C ⎟ β g 1 IC = ⎝ F⎠= = m βF VT βF dVBE (assuming βF = const.) ⎡ VBE ⎤ dIC VCE ⎞ ⎥ IC 1 d ⎢ VT ⎛ = go = = ISe ⎜1+ ⎟ ≅ r0 dVCE dVCE ⎢ VA ⎠ ⎥ VA ⎝ ⎣ ⎦ B. Murmann EE214 Winter 2010-11 – Chapter 2 21 Intrinsic Gain gmro ≅ IC VA VA ⋅ = VT IC VT VT ≅ 26mV (at room temperature) In the EE214 technology, technology the SiGe npn device has VA = 90V, 90V thus gmro ≅ 90V = 3460 26mV Much larger than the intrinsic gain of typical MOSFET devices B. Murmann EE214 Winter 2010-11 – Chapter 2 22 Outline – Model Extensions and Technology Complete picture of BJT operating regions Dependence p of βF on operating p g conditions Device capacitances and resistances Technology – Junction J ti isolated i l t d – Oxide isolated with polysilicon emitter – Heterojunction bipolar (SiGe base) – BiCMOS – Complementary bipolar B. Murmann EE214 Winter 2010-11 – Chapter 2 23 BJT Operating Regions Discussed so far BE = forward biased CE = reverse biased Text, p. 17 B. Murmann EE214 Winter 2010-11 – Chapter 2 24 Carrier Concentrations in Saturation Text, p. 16 Base-Collector junction is forward biased np(WB), and therefore also IC, strongly depend on VBC, VCE VCE(sat) is the voltage at which the devices enters saturation – The difference between the two junction voltages, small ~0.05…0.3V 0.05…0.3V B. Murmann EE214 Winter 2010-11 – Chapter 2 25 Gummel Plot A Gummel G l plot l t iis a semi-log i l plot l t off IC and d IB versus VBE (linear (li scale) l ) It reveals the regions for which high βF is maintained (region II below) What happens pp in regions g I and III? Text, p. 24 B. Murmann EE214 Winter 2010-11 – Chapter 2 26 βF Fall-Off Region III (high current density) – Injected electron charge in base region nears the level of doping (“high level injection”) – For this case, it can be shown that the injected carrier concentration rises with a smaller exponent (cut in half) and therefore IC = ISe 1 VBE 2 VT Region I (low current density) – There exists excess base current due to (unwanted) recombination in the depletion layer of the base-emitter junction – This current becomes significant at low current densities and sets a minimum for IB B. Murmann EE214 Winter 2010-11 – Chapter 2 27 Current Profile of a Forward Biased Diode Revisited Extra current due to recombination (small) Total current Recombination B. Murmann EE214 Winter 2010-11 – Chapter 2 28 βF vs. IC and Temperature Text, p. 24 ≅ 7000 ppm / °C B. Murmann EE214 Winter 2010-11 – Chapter 2 29 Junction Isolated npn Transistor Text, p. 97 B. Murmann EE214 Winter 2010-11 – Chapter 2 30 Device Capacitances and Resistances Big mess! First focus on intrinsic elements B. Murmann EE214 Winter 2010-11 – Chapter 2 31 Charge Storage IIn the th intrinsic i t i i transistor, t i t charge h is i stored t d in i the th jjunction ti capacitances, it Cje and Cjc = Cμ, and as minority carriers in the base and emitter Both minority carrier charge injected into the base and into the emitter, are proportional ti l to t exp(V (VBE/VT) – But the charge in the base is much larger, as discussed previously Base terminal must supply charge for neutrality Voltage dependent charge Æ capacitance (Cb) ΔVBE causes change in injected charge Text, p. 26 B. Murmann EE214 Winter 2010-11 – Chapter 2 32 Base Charging Capacitance Cb ∂Qe ∂Qe ∂IC = = τFgm ∂VBE ∂IC ∂VBE τF = ∂Qe ∂ ⎛1 ⎞ = np (0)WBqA ⎟ ∂IC ∂IC ⎜⎝ 2 ⎠ τF = ∂ ⎛ 1 WB2 ⎞ 1 WB2 IC ⎟ = ⎜ ∂IC ⎜⎝ 2 Dn ⎟⎠ 2 Dn IC = qADnnp (0) WB τF is called the base transit time (in forward direction) Typical values for high-speed transistors are on the order of 1…100ps B. Murmann EE214 Winter 2010-11 – Chapter 2 33 Junction Capacitance Cj = C j0 2C j0 n ⎛ VD ⎞ ⎜1− ⎟ ⎝ ψ0 ⎠ Text, p. 6 B. Murmann EE214 Winter 2010-11 – Chapter 2 34 Small-Signal Model with Intrinsic Capacitances Cμ B v1 – rπ C gmv1 Cπ ro E Cπ = Cb + C je = Cb + 2C je0 Cμ = C jc = B. Murmann C jc0 n ⎛ VCB ⎞ ⎜1+ ⎟ ψ 0c ⎠ ⎝ EE214 Winter 2010-11 – Chapter 2 35 Model with Additional Parasitics Neglect Text, p. 32 Range of numbers re ~1-3Ω Values at high end of these ranges may have large rb ~ 50-500Ω 50 500Ω impact on performance Æ Try to minimize through rc ~ 20-500Ω advanced processing & technology CCS ~ 3-200fF B. Murmann EE214 Winter 2010-11 – Chapter 2 36 BJT in Advanced Technology Text, p. 107 Oxide isolated Self-aligned structure (base and emitter align automatically) Very thin base (~100nm or less) through ion implantation Reduced breakdown voltages compared to more traditional structures B. Murmann EE214 Winter 2010-11 – Chapter 2 37 SiGe Heterojunction Bipolar Technology A heterojunction is a pn junction formed with different materials for the n and p regions Germanium is added to the base of a silicon bipolar transistor to create a heterojunction bipolar transistor (HBT) – Base formed by growing a thin epitaxial layer of SiGe – Results in a lower bandgap (and higher intrinsic carrier concentration) in the base than emitter In “band diagram speak” the bandgap mismatch increases the barrier to the injection of holes (in an npn transistor) from the base into the emitter One way to enumerate the benefits of a SiGe base is to look at the current gain expression B. Murmann EE214 Winter 2010-11 – Chapter 2 38 HBT Current Gain Intrinsic carrier concentration in the SiGe base (niB) is larger than intrinsic carrier concentration in the Si emitter (niE) qADnnp0 βF = WB qADnnp0 2 1 np0 WBqA qADpniE + τb 2 LpND ≅ WB 2 qADpniE = 2 qADnniB WBNA 2 qADpniE LpND = DnNDLp ⋅ 2 niB 2 DpNA WB niE LpND Added degree of freedom for HBT Base doping (NA) can be increased while maintaining same βF − Can reduce base width without affecting rb − Larger ro due to decrease in base width modulation B. Murmann EE214 Winter 2010-11 – Chapter 2 39 Device Parameter Comparison SiGe npn HBT Transistor with 0.7μm2 = 0.22μm x 3.2μm Emitter Area 300 2 90 3.2x10-17A 1pA 2.0V 5 5V 5.5V 3.3V 0.56ps 10ps 25Ω 60 60Ω 2.5Ω 6.26fF 0.8V 0.4 3.42fF 0.6V 0.33 3.0fF 0 6V 0.6V 0.33 B. Murmann EE214 Winter 2010-11 – Chapter 2 3x smaller device 5x bigger IS 18x smaller τF 16x smaller rb Oxide isolation vs. Junction isolation 40 BiCMOS Technology CMOS Text, p. 154 Older BJTs used poly Si as “diffusion source” for emitter doping. doping Advanced (state-of-the-art) BJTs use epitaxial growth of both the SiGe base and Si emitter regions B. Murmann EE214 Winter 2010-11 – Chapter 2 41 Advanced Complementary Bipolar Technology [Texas Instruments] B. Murmann EE214 Winter 2010-11 – Chapter 2 42 Cross Section [Texas Instruments] B. Murmann EE214 Winter 2010-11 – Chapter 2 43 Figures of Merit for BJTs Product of current gain and Early voltage, β·VA Product of transit frequency q y and breakdown voltage, g , fT·BVCEO Maximum frequency of oscillation, fmax – More in EE314 T Transit it (or ( transition) t iti ) frequency, f fT – Formally defined as the frequency for which the current gain of the device falls to unity – Important to keep in mind that the basic device model may fall apart altogether at this frequency • Lumped device models tend to be OK up to ~fT/5 – Therefore, Therefore fT should be viewed as an extrapolated parameter parameter, or simply as a proxy for device transconductance per capacitance B. Murmann EE214 Winter 2010-11 – Chapter 2 44 Transit Frequency Calculation (1) Ignore for simplicity Text, p. 35 v1 = (AC circuit; DC biasing not shown) 1 + rπs Cπ + Cμ io gmrπ = ≅ gmrπ = βF ii 1 + rπ jω Cπ + Cμ ( ) io gm ≅ ii jω Cπ + Cμ ( ( rπ for ω << io = gmv1 i )i ( 1 rπ Cπ + Cμ ) = ωβ for ω >> ωβ ) B. Murmann EE214 Winter 2010-11 – Chapter 2 45 Transit Frequency Calculation (2) |io/ii| ( gm ω Cπ + Cμ ) (asymptote) ≅ βF Note that rπ “matters” only for frequencies up to ωβ = ωT/βF Text, p. 36 1= ωT Cπ + Cμ τT = B. Murmann ( gm ) ⇒ ωT = gm Cπ + Cμ C je Cμ C je Cμ C 1 = b+ + = τF + + gm gm ωT gm gm gm EE214 Winter 2010-11 – Chapter 2 46 fT versus IC plot “peak fT” Cje and Cμ dominate High level injection, τF increases Text, p. 37 gm = IC/VT increases The particular current value at which fT is maximized depends on the particular parameters of a technology and the emitter area of the BJT B. Murmann EE214 Winter 2010-11 – Chapter 2 47 EE214 Technology Assumed to be similar to a 0 0.18 18-μm μm BiCMOS technology featuring a high-performance SiGe npn device – VCC = 2.5V (BJT), VDD=1.8V (MOS) See e e.g. g – Wada et al., BCTM 2002 – Joseph et al., BCTM 2001 – IBM 7HP documentation • https://www-01.ibm.com/chips/techlib/techlib.nsf/products/BiCMOS_7HP) NPN PMOS/NMOS Poly resistor http://fuji.stanford.edu/events/spring01/slides/harameSlides.pdf B. Murmann EE214 Winter 2010-11 – Chapter 2 48 Cross Section of npn Device B. Murmann EE214 Winter 2010-11 – Chapter 2 49 EE214 npn Unit Device A technology typically comes with an optimized layout for a unit device of a certain size AE = 0.22μm x 3.2μm Great care is then taken to extract a Spice model for this particular layout using measured data Spice model (usr/class/ee214/hspice/ee214_hspice.sp) E .model npn214 npn + level=1 tref=25 is=.032f bf=300 br=2 vaf=90 B + cje=6.26f vje=.8 mje=.4 cjc=3.42f vjc=.6 mjc=.33 + re=2 re=2.5 5 rb=25 rc=60 tf=563f tr=10p C + xtf=200 itf=80m ikf=12m ikr=10.5m nkf=0.9 Instantiation in a circuit netlist B. Murmann * C q1 n1 n2 n3 B E npn214 EE214 Winter 2010-11 – Chapter 2 50 BJT Model Parameters For more info consult the HSpice documentation under /afs/ir.stanford.edu/class/ee/synopsys/B-2008.09-SP1/hspice/docs_help PDF files: home.pdf hspice_cmdref.pdf hspice_integ.pdf hspice_relnote.pdf hspice_sa.pdf hspice_devmod.pdf hspice_mosmod.pdf hspice_rf.pdf hspice_si.pdf B. Murmann EE214 Winter 2010-11 – Chapter 2 51 Adding Multiple Devices in Parallel For the unit device, there exists a practical upper bound for the collector current – Due D tto th the onsett off high hi h llevell iinjection j ti This means that the unit device can only deliver a certain maximum gm E E E B B B If more gm is needed, “m” unit devices can be connected in parallel C C C Instantiation in a circuit netlist ((m=3)) B. Murmann * C q1 n1 n2 n3 B E EE214 Winter 2010-11 – Chapter 2 npn214 3 52 npn Unit Device Characterization * ee214 npn device characterization * q1 Vc ib C c c 0 B E b 0 0 b .op .dc ib .probe .probe .probe b .probe .probe .probe npn214 1.25 1u dec 10 10f 100u ib(q1) ic(q1) ie(q1) gm = par('gm(q1)') go = par('g0(q1)') (' 0( 1)') cpi = par('cap_be(q1)') cmu = par('cap_ibc(q1)') beta = par('beta(q1)') CC .options dccap post brief .inc '/usr/class/ee214/hspice/ee214_hspice.sp' .end B. Murmann EE214 Winter 2010-11 – Chapter 2 53 DC Operating Point Output **** bipolar junction transistors element 0:q1 model 0:npn214 ib 999.9996n ic 288.5105u vbe 803.4402m vce 1.2500 vbc -446.5598m vs -1.2327 power 361.4415u betad 288.5106 gm 10.2746m rpi 26 8737k 26.8737k rx 25.0000 ro 313.4350k cpi 14.6086f cmu 2 8621f 2.8621f cbx 0. ccs 0. betaac 276.1163 g ft 93.5999g B. Murmann EE214 Winter 2010-11 – Chapter 2 54 Gummel Plot B. Murmann EE214 Winter 2010-11 – Chapter 2 55 Transit Frequency fT [GH Hz] 150 100 50 0 10 -4 10 -3 10 -2 IC [A] B. Murmann EE214 Winter 2010-11 – Chapter 2 56 Intrinsic Gain ee215 npn 4000 3500 gm /go [GHz]] 3000 2500 2000 1500 1000 500 0 10 -8 10 -6 10 -4 10 -2 IC [A] B. Murmann EE214 Winter 2010-11 – Chapter 2 57 gm/IC Important to realize that gm will not be exactly equal to IC/VT at high currents B. Murmann EE214 Winter 2010-11 – Chapter 2 58 I-V Curves NPN (1x, AE=0.7μm2, I B=0.2, 0.4, ..., 1μA 300 800 NMOS 2/0.18, VGS=0.6, 0.8, ..., 1.4V 700 250 600 500 I D [μA] I C [μA] 200 150 400 300 100 200 50 0 0 B. Murmann 100 0.5 1 1.5 VCE [V] 2 2.5 0 0 EE214 Winter 2010-11 – Chapter 2 0.5 1 VDS [V] 1.5 59 Passive Components (1) Joseph et al., BCTM 2001 B. Murmann EE214 Winter 2010-11 – Chapter 2 60 Passive Components (2) Diffusion Resistor MIM Capacitor Text, p. 116 B. Murmann EE214 Winter 2010-11 – Chapter 2 61 Chapter 3 MOS Transistor Modeling Gm/ID-based based Design B. Murmann Stanford University Reading Material: Sections 1.5, 1.6, 1.7, 1.8 Basic MOSFET Operation (NMOS) Text,, p.41 p How to calculate drain current (ID) current as a function of VGS, VDS? B. Murmann EE214 Winter 2010-11 – Chapter 3 2 Simplifying Assumptions 1)) Current is controlled byy the mobile charge g in the channel. This is a very y good approximation. 2) "Gradual Channel Assumption" - The vertical field sets channel charge, so we can approximate the available mobile charge through the voltage diff difference b between t th the gate t and d th the channel h l 3) The last and worst assumption (we will fix it later) is that the carrier velocity is proportional to lateral field (ν = μE). This is equivalent to Ohm's law: velocity (current) is proportional to E-field E field (voltage) B. Murmann EE214 Winter 2010-11 – Chapter 3 3 Derivation of First Order IV Characteristics (1) Qn (y) = Cox [ VGS − V(y) − Vt ] ID = Qn ⋅ v ⋅ W v = μ ⋅E ID = Cox [ VGS − V(y) − Vt ] ⋅ μ ⋅ E ⋅ W ID = μCox B. Murmann VDS ⎤ W⎡ ⎢( VGS − Vt ) − ⎥ ⋅ VDS L ⎣ 2 ⎦ EE214 Winter 2010-11 – Chapter 3 4 Pinch-Off – VGS + + VDS – N Qn(y), V(y) N Voltage at the end of channel is fixed at VGS-V Vt y y=0 y=L Effective voltage across channel is VGS-Vt – At the point where channel charge goes to zero, there is a high lateral field that sweeps the carriers to the drain • Recall that electrons are minority carriers in the p-region of a pn j junction; ti th they are b being i sweptt ttoward d th the n-region i – The extra drain voltage drops across depletion region To first order, the current becomes independent of VDS B. Murmann EE214 Winter 2010-11 – Chapter 3 5 Plot of Output Characteristic Saturation Region ID Triode Region VGS-V Vt VDS Triode Region: ID = μCox VDS ⎤ W⎡ ⎢( VGS − Vt ) − ⎥ ⋅ VDS L ⎣ 2 ⎦ Saturation Region: ID = μCox (V − V ) W⎡ ( VGS − Vt ) − GS t ⎤⎥ ⋅ (VGS − Vt ) ⎢ L ⎣ 2 ⎦ = B. Murmann 1 W (VGS − Vt )2 μCox 2 L EE214 Winter 2010-11 – Chapter 3 6 ID Plot of Transfer Characteristic (in Saturation) Vt VGS VOV gm = dID W W = μCox ( VGS − Vt ) = μCox VOV dVGS L L = 2IDμCox B. Murmann 2I W = D L VOV EE214 Winter 2010-11 – Chapter 3 7 Output Characteristic with “Channel Length Modulation” Triode Region Saturation Region ID go= dID/dVDS ≠ 0 VGS-Vt VDS go = = B. Murmann dID d = dVDS dVDS W ⎡1 ⎤ 2 ⎢ 2 μCox L (VGS − Vt ) (1 + λVds )⎥ ⎣ ⎦ λID 1 W μCox (VGS − Vt )2 ⋅ λ = ≅ λID 2 L 1 + λVDS EE214 Winter 2010-11 – Chapter 3 8 Capacitances Text, p. 54 B. Murmann EE214 Winter 2010-11 – Chapter 3 9 Gate Capacitance Summary B. Murmann Subthreshold Triode Saturation Cgs Col ½ WLCox+ Col Cgd Col ½ WLCox+Col Col Cgb ⎛ 1 1 ⎞ + ⎜⎜ ⎟⎟ ⎝ C js WLCox ⎠ 0 0 2/ 3 WLCox + Col −1 EE214 Winter 2010-11 – Chapter 3 10 Capacitance Equations and Parameters Parameter Cdb = ε Cox = ox t ox Col = WCol' AD ⋅ CJ MJ VDB ⎞ ⎛ ⎜ 1 + PB ⎟ ⎝ ⎠ + PD ⋅ CJSW MJSW VDB ⎞ ⎛ ⎜ 1 + PB ⎟ ⎝ ⎠ AD = WL diff PD = W + 2L diff NMOS PMOS Cox 8.42 fF/μm2 8.42 fF/μm2 C’ol C 0 491 fF/μm 0.491 0 657 fF/μm 0.657 CJ 0.965 fF/μm2 1.19 fF/μm2 CJSW 0 233 fF/μm 0.233 0 192 fF/μm 0.192 PB 0.8 V 0.8 V MJ 0.38 0.40 MJSW 0.13 0.33 0.64 μm 0.64 μm LDIF B. Murmann EE 214 Technology (0.18μm) EE214 Winter 2010-11 – Chapter 3 11 Complete Small-Signal Model gd gs gs gb sb m gs o db 2 bsub (Cj0 = 0.2 fF/μm , “dwell” model) Cgg Cgs + Cgb + Cgd B. Murmann Cdd Cdb + Cgd EE214 Winter 2010-11 – Chapter 3 12 What are μCox (“KP”) and λ (“LAMBDA”) for our Technology? .MODEL MODEL nmos214 nmos +acm hdif = 0.32e-6 LEVEL = 49 +VERSION = 3.1 TNOM = 27 TOX = 4.1E-9 +XJ = 1E-7 = 3 NCH = 2.3549E17 VTH0 = 0.3618397 +K1 = 0.5916053 K2 = 3.225139E-3 +K3B = 2.3938862 W0 = 1E-7 NLX = 1.776268E-7 +DVT0W = 0 DVT1W = 0 DVT2W = 0 +DVT0 = 1.3127368 DVT1 = 0.3876801 DVT2 = 0.0238708 +U0 = 256.74093 UA = -1.585658E-9 UB = 2.528203E-18 +UC = 5.182125E-11 VSAT = 1.003268E5 A0 = 1.981392 +AGS = 0.4347252 B0 = 4.989266E-7 B1 +KETA = -9.888408E-3 A1 = 6.164533E-4 A2 = 0.9388917 +RDSW = 128.705483 PRWG = 0.5 PRWB = -0.2 +WR = 1 WINT = 0 LINT = 1.617316E-8 +XL = 0 XW = -1E-8 DWG = -5.383413E-9 +DWB = 9.111767E-9 VOFF = -0.0854824 NFACTOR = 2.2420572 +CIT = 0 CDSC = 2.4E-4 CDSCD = 0 +CDSCB = 0 ETA0 = 2.981159E-3 ETAB = 9.289544E-6 +DSUB = 0.0159753 PCLM = 0.7245546 PDIBLC1 = 0.1568183 K3 = 1E-3 = 5E-6 +PDIBLC2 = 2.543351E-3 PDIBLCB = -0.1 DROUT +PSCBE1 = 8E10 PSCBE2 = 1.876443E-9 PVAG = 7.200284E-3 +DELTA = 0.01 RSH = 6.6 MOBMOD = 1 +PRT = 0 UTE = -1.5 KT1 = -0.11 +KT1L = 0 KT2 = 0.022 UA1 = 4.31E-9 +UB1 = -7.61E-18 UC1 = -5.6E-11 AT = 3.3E4 +WL = 0 WLN = 1 WW = 0 +WWN = 1 WWL = 0 LL = 0 +LLN = 1 LW = 0 LWN = 1 +LWL = 0 CAPMOD = 2 XPART = 1 +CGDO = 4.91E-10 CGSO = 4.91E-10 CGBO = 1E-12 +CJ = 9.652028E-4 PB = 0.8 MJ = 0.3836899 +CJSW = 2.326465E-10 PBSW = 0.8 MJSW = 0.1253131 +CJSWG = 3.3E-10 PBSWG = 0.8 MJSWG = 0.1253131 +CF = 0 PVTH0 = -7.714081E-4 PRDSW = -2.5827257 +PK2 = 9.619963E-4 WKETA = -1.060423E-4 LKETA = -5.373522E-3 +PU0 = 4.5760891 4 5760891 PUA = 1.469028E 1 469028E-14 14 PUB = 1.783193E 1 783193E-23 23 +PVSAT = 1.19774E3 PETA0 = 9.968409E-5 PKETA = -2.51194E-3 +nlev = 3 kf = 0.5e-25 B. Murmann The HSpice p model for an NMOS device in our technology is shown to the left BSIM 3v3 model 110 parameters KP and LAMBDA nowhere to be found found… = 0.7445011 EE214 Winter 2010-11 – Chapter 3 13 An Attempt to Extract μCox Bias MOSFET at constant VDS>VOV, sweep VGS and plot μCox estimate 300 NMOS 5/0.18 NMOS 20/0.72 μCox 2ID = W 2 V L OV 2 μnCox [μA/V ] 250 200 150 100 50 0 0 0.1 0.2 V OV 0.3 [V] 0.4 0.5 The extracted μCox depends on L and VOV and cannot be viewed as a constant parameter B. Murmann EE214 Winter 2010-11 – Chapter 3 14 Questions Which physical effects explain the large deviation from the basic square law model? How can we design with such a device? – Is there another “simple” equation that describes its behavior? We will approach the above two questions by performing a systematic, simulation-based device characterization – And discuss the relevant p physical y p phenomena that explain p the observed behavior As a basis for this characterization, we consider three basic figures of merit that relate directlyy to circuit design g B. Murmann EE214 Winter 2010-11 – Chapter 3 15 Figures of Merit for Device Characterization Square Law Transconductance efficiency – Want W t large l gm, for f as little littl currentt as possible gm ID = 2 VOV Transit frequency – Want large g gm, without large g Cgg gm Cgg ≅ 3 μVOV 2 L2 Intrinsic gain – Want large gm, but no go gm go ≅ 2 λVOV B. Murmann EE214 Winter 2010-11 – Chapter 3 16 Device Characterization * NMOS characterization .param .param vds vgs mn .op .dc gs .probe .probe b .probe .probe gs=0.7 dd=1.8 d 0 g 0 d g 0 0 dc 'dd/2' dc 'gs' nmos214 L=0.18um W=5um DD 0.2V 1V 10mV ov = par('gs-vth(mn)') gm_id id = par('gmo(mn)/i(mn)') (' ( )/i( )') ft = par('1/6.28*gmo(mn)/cggbo(mn)') gm_gds = par('gmo(mn)/gdso(mn)') .options options post brief dccap .inc /usr/class/ee214/hspice/ee214_hspice.sp .end B. Murmann EE214 Winter 2010-11 – Chapter 3 17 gm/ID Plot 40 35 0.18um NMOS 2/VOV gm/I D [S/A A] 30 BJT (q/kT) 25 20 15 10 5 0 -0.2 B. Murmann -0.1 0 0.1 0.2 VOV [V] EE214 Winter 2010-11 – Chapter 3 0.3 0.4 0.5 18 Observations Square law prediction is fairly close for VOV > 150mV Unfortunatelyy gm/ID does not approach pp infinity y for VOV → 0 It also seems that we cannot do better than a BJT, even though the square law equation would predict this for 0 < VOV < 2kT/q ≅ 52mV For further analysis, analysis it helps to identify three distinct operating regions – Strong inversion: VOV > 150mV • Deviations due to short channel effects – Subthreshold: VOV < 0 • Behavior similar to a BJT, gm/ID nearly constant – Moderate Inversion: 0 < VOV < 150mV • Transition region, region an interesting mi mix of the abo above e B. Murmann EE214 Winter 2010-11 – Chapter 3 19 Subthreshold Operation A plot of the device current in our previous simulation 1 10 0 0.8 10 -1 0.6 10 -2 10 -3 10 -4 I D [mA A] I D [mA A] 0.4 0.2 0 -0.5 -5 0 0.5 V 1 10 -0.5 [V] 0 0.5 V VOV [V] 1 [V] VOV [V] Questions – What determines the current when VOV< 0, i.e. VGS< Vt? – What Wh t iis th the d definition fi iti off Vt? B. Murmann EE214 Winter 2010-11 – Chapter 3 20 Definition of Vt Vt is defined as the VGS at which the number of electrons at the surface equals the number of doping atoms Seems somewhat arbitrary, but makes sense in terms of surface charge control B. Murmann EE214 Winter 2010-11 – Chapter 3 21 Mobile Charge versus VOV 6.E-07 Ch harge [C] 5.E-07 Fixed Charge Mobile Charge T t l Charge Total Ch 4.E-07 3.E-07 2.E-07 1.E-07 0.E+00 -1.0 -0.5 0.0 0.5 1.0 VOV [V] Around VGS=Vt (VOV=0), the relationship between mobile charge in the channel and gate voltage becomes linear (Qn ~ CoxVOV) – Exactly what we assumed to derive the long channel model B. Murmann EE214 Winter 2010-11 – Chapter 3 22 Mobile Charge on a Log Scale O On a log l scale, l we see th thatt there th are mobile bil charges h b f before we reach h the threshold voltage – Fundamental result of solid-state physics, not short channels 1.E-06 1.E-07 Mobile Charge [C] 1 E-08 1.E 08 1.E-09 1.E-10 1.E-11 1.E-12 1.E-13 1.E-14 1.E-15 1.E-16 -1.00 -0.50 0.00 0.50 1.00 VOV [V] B. Murmann EE214 Winter 2010-11 – Chapter 3 23 BJT Similarity We have – An NPN sandwich, sandwich mobile minority carriers in the P region This is a BJT! – Except that the base potential is here controlled through a capacitive divider,, and not directlyy byy an electrode B. Murmann EE214 Winter 2010-11 – Chapter 3 24 Subthreshold Current • We know that for a BJT • kT q For the MOSFET in subthreshold we have VBE IC = IS ⋅ e VT VT = ID = I0 ⋅ e • VGS − Vt nV VT n is given by the capacitive divider n= C js + Cox Cox = 1+ C js Cox where Cjs is the depletion layer capacitance • In the EE214 technology n ≅ 1.5 B. Murmann EE214 Winter 2010-11 – Chapter 3 25 Comparison – NMOS versus NPN Vt B. Murmann EE214 Winter 2010-11 – Chapter 3 26 Subthreshold Transconductance gm = gm 1 = ID nVT dID 1I = D dVGS n VT Similar to BJT, but unfortunately n (≅1.5) times lower 40 35 30 gm/I D [S/A A] 0.18um NMOS 2/VOV ~1.5x BJT (q/kT) 25 20 15 10 5 0 -0.2 02 -0.1 01 0 B. Murmann 0.1 0 1 02 0.2 VOV [V] 03 0.3 04 0.4 05 0.5 EE214 Winter 2010-11 – Chapter 3 27 Moderate Inversion IIn the th transition t iti region i between b t subthreshold bth h ld and d strong t inversion, i i we have two different current mechanisms Drift (MOS) ν = μE Diffusion ((BJT)) ν = D dn kT dn = μ dx q dx Both current components are always present – Neither one clearly dominates in moderate inversion Can show that ratio of drift/diffusion current ~(VGS-Vt)/(kT/q) – MOS equation becomes dominant at several kT/q B. Murmann EE214 Winter 2010-11 – Chapter 3 28 Re-cap Subthreshold Operation ? Transition to Strong Inversion What causes the discrepancy between 2/VOV and 0.18μm NMOS in strong inversion? B. Murmann EE214 Winter 2010-11 – Chapter 3 29 Short Channel Effects Velocity saturation due to high lateral field Mobility y degradation g due to high g vertical field Vt dependence on channel length and width Vt = f(VDS) ro = f(VDS) … We will limit the discussion in EE214 to the first two aspects of the above list, with a focus on qualitative understanding B. Murmann EE214 Winter 2010-11 – Chapter 3 30 Velocity Saturation (1) IIn the th derivation d i ti off the th square llaw model, d l it iis assumed d th thatt th the carrier i velocity is proportional to the lateral E-field, v=μE Unfortunately, the speed of carriers in silicon is limited (vscl ≅ 105 m/s) – At very high fields (high voltage drop across the conductive channel), the carrier velocity saturates Text, p. 60 (approximation) ν d (E) ≅ μE E 1+ Ec ≅ μEc = v scl = B. Murmann EE214 Winter 2010-11 – Chapter 3 v scl 2 for E >> Ec for E = Ec 31 Velocity Saturation (2) It is i important i t t to t distinguish di ti i h various i regions i iin th the above b plot l t – Low field, the long channel equations still hold – Moderate field, the long channel equations become somewhat inaccurate – Very high field across the conducting channel – the velocity saturates completely and becomes essentially constant (vscl) T To gett some feel f l for f latter l tt two t cases, let's l t' first fi t estimate ti t the th E field fi ld using i simple long channel physics In saturation, the lateral field across the channel is E = B. Murmann VOV L e.g. 200mV V = 1.11⋅ 106 0.18μm m EE214 Winter 2010-11 – Chapter 3 32 Field Estimates In our 0.18μm technology, we have for an NMOS device v = scl μ Ec Therefore m s = 6.7 ⋅ 106 V ≅ cm2 m 150 Vs 105 V 1.11⋅ 106 E m = ≅ 0.16 V Ec 6.7 ⋅ 106 m This means that for VOV on the order of 0.2V, the carrier velocity is p is relatively y small somewhat reduced,, but the impairment The situation changes when much larger VOV are applied, e.g. as the case in digital circuits B. Murmann EE214 Winter 2010-11 – Chapter 3 33 Short Channel ID Equation A simple equation that captures the moderate deviation from the long channel drain current can be written as (see text, p. 62) ID ≅ ≅ 1 W 2 1 μCox VOV ⋅ 2 L ⎛ VOV ⎞ ⎜1+ ⎟ ⎝ Ec L ⎠ E L ⋅ VOV 1 W VOV ⋅ c μCox 2 L (EcL + VOV ) Think of this as a “parallel combination" V ⋅0 0.18 18μm = 1 1.2V 2V m Minimum-length Minimum length NMOS: Ec L = 6 6.7 7 ⋅ 106 Minimum-length PMOS: EcL = 16.75 ⋅ 106 B. Murmann V ⋅ 0.18μm = 3V m EE214 Winter 2010-11 – Chapter 3 34 Modified gm/ID Expression Assuming VOV << EcL, we can show that (see text, pp. 63-64) gm 2 1 ≅ ⋅ ID VOV ⎛ VOV ⎞ ⎜1+ ⎟ ⎝ Ec L ⎠ E.g. E for f an NMOS device d i with ith VOV=200mV 200 V gm 2 1 2 ≅ ⋅ = ⋅ 0.86 ID VOV ⎛ 0.2 ⎞ VOV 1 + ⎜ ⎟ ⎝ 1.2 ⎠ Means that the square q law model in strong g inversion ((at VOV ≅ 200mV)) should be off by about 15% This prediction agrees well with the simulation data B. Murmann EE214 Winter 2010-11 – Chapter 3 35 Mobility Degradation due to Vertical Field In MOS technology, the oxide thickness has been continuously scaled down with feature size – ~6.5nm in 0.35μm, ~4nm in 0.18μm, ~1.8nm in 90nm CMOS As a result, the vertical electric field in the device increases and tries to pull the carriers closer to the "dirty" silicon surface – Imperfections impede movement and thus mobility This effect can be included by replacing the mobility term with an "effective mobility" μ eff ≅ μ (1 + θVOV ) θ = 0.1...0.4 1 V Yet another "fudge factor" – Possible to lump with EcL parameter, if desired B. Murmann EE214 Winter 2010-11 – Chapter 3 36 Transit Frequency Plot fT = B. Murmann 1 gm 2π Cgg EE214 Winter 2010-11 – Chapter 3 37 Observations - fT Again the square-law model doesn't do a very good job – Large fT discrepancy in subthreshold operation and in strong inversion (large VOV) The reasons for these discrepancies are exactly the same as the ones we came across when looking at gm/ID – Bipolar action in subthreshold operation and moderate inversion – Short channel effects at large VOV • Less gm, hence lower gm/Cgg Same conclusion: we won't be able to make good predictions with a simple square law relationship B. Murmann EE214 Winter 2010-11 – Chapter 3 38 gm/ID· fT Plot Square Law: gm 1 3μ ⋅ fT ≅ ID 2π L2 Sweet spot (?) Square law predicts too much gm/ID B. Murmann Short channel effects EE214 Winter 2010-11 – Chapter 3 39 Intrinsic Gain Plot Impossible to approximate with the “λ” model equation! B. Murmann EE214 Winter 2010-11 – Chapter 3 40 Gradual Onset of 1/gds VDS = VOV B. Murmann EE214 Winter 2010-11 – Chapter 3 41 Gradual Onset of 1/gds (Zoom) VDS = VOV B. Murmann VDS = 2/(gm/ID) EE214 Winter 2010-11 – Chapter 3 42 “VDSsat” Estimate Based on gm/ID “VDSsat” defined (arbitrarily) as VDS at which 1/g gds is equal q to ½ of the value at VDS = VDD/2 = 0.9V ≅4kT/q V* = 2/(gm/ID) is a reasonable estimate of “V VDSsat” B. Murmann EE214 Winter 2010-11 – Chapter 3 43 Observations – Intrinsic Gain Device shows a rather gradual transition from triode to saturation – Square law predicts an abrupt change from small to large intrinsic gain at VDS = VOV – V* = 2/(gm/ID) provides a reasonable estimate for the minimum VDS that is needed to extract gain from a device • Typically T i ll wantt tto stay t att lleastt 100 100mV V above b thi this value l in i practical ti l designs The physics that govern the behavior of ro=1/gds are complex – Channel length modulation – Drain induced barrier lowering (DIBL) – Substrate current induced body effect (SCBE) • Not present in all technologies and/or PMOS devices If you are interested in more details, please refer to EE316 or similar B. Murmann EE214 Winter 2010-11 – Chapter 3 44 The Challenge Square-law model is inadequate for design in fine-line CMOS – But simulation models (BSIM, PSP, …) are too complex for handcalculations This issue tends to drive many designers toward a “spice monkey” design methodology – No hand calculations, iterate in spice until the circuit “somehow” meets t the th specifications ifi ti – Typically results in sub-optimal designs Our goal – Maintain a systematic design methodology in absence of a set of compact MOSFET equations Strategy – Design using look-up tables or charts [[Courtesyy Isaac Martinez]] B. Murmann EE214 Winter 2010-11 – Chapter 3 45 The Problem B. Murmann EE214 Winter 2010-11 – Chapter 3 46 The Solution Use pre-computed spice data in hand calculations B. Murmann EE214 Winter 2010-11 – Chapter 3 47 Technology Characterization for Design Pl Plott the th following f ll i parameters t ffor a reasonable bl range off gm/ID and d channel lengths – Transit frequency (fT) – Intrinsic gain (gm/gds) – Current density (ID/W) In addition, may want to tabulate relative estimates of extrinsic capacitances it – Cgd/Cgg and Cdd/Cgg Parameters are (to first order) independent of device width – Enables "normalized design" and re-use of charts – Somewhat similar to filter design procedure using normalized coefficient tables Do hand calculations using the generated technology data – Can use Matlab functions to do table-look-up on pre-computed data B. Murmann EE214 Winter 2010-11 – Chapter 3 48 NMOS Simulation Data B. Murmann EE214 Winter 2010-11 – Chapter 3 49 PMOS Simulation Data B. Murmann EE214 Winter 2010-11 – Chapter 3 50 Transit Frequency Chart L=0.18um L=0.5um B. Murmann EE214 Winter 2010-11 – Chapter 3 51 Intrinsic Gain Chart L=0.5um L 0 18 L=0.18um B. Murmann EE214 Winter 2010-11 – Chapter 3 52 Current Density Chart L=0.18um L 05 L=0.5um B. Murmann EE214 Winter 2010-11 – Chapter 3 53 Lookup Functions in Matlab % Set up path and load simulation data (for VDS=0.9V) addpath('/usr/class/ee214/matlab'); load techchar.mat; % Lookup fT for NMOS, L=0.18um, at gm/ID=10S/A lookup_ft(tech, 'n', 0.18e-6, 10) ans = 2.2777e+10 % Lookup gm/ID for NMOS, L=0.18um, at fT=20GHz lookup_gmid(tech, 'n', 0.18e-6, 20e9) ans = 11.5367 % Lookup ID/W for NMOS, L=0.18um, at gm/ID=10S/A lookup_idw(tech, 'n', 0.18e-6, 10) ans = 29.3281 B. Murmann EE214 Winter 2010-11 – Chapter 3 54 VDS Dependence B. Murmann EE214 Winter 2010-11 – Chapter 3 VDS dependence is relatively weak Typically yp y OK to work with data generated for VDD/2 55 Extrinsic Capacitances (1) NMOS, L=0.18um 1 Cdd/Cgg Cgd/Cgg 0.8 0.60 06 0.6 Again, usually OK to work with estimates taken at VDD/2 0.4 0 24 0.24 0.2 0 0 B. Murmann 0.5 1 VDS [V] 1.5 EE214 Winter 2010-11 – Chapter 3 56 Extrinsic Capacitances (2) NMOS, gm/I D=10S/A, VDS=0.9V 0.8 Cgd/Cgg 07 0.7 Cdd/Cgg 0.6 0.5 0.4 0.3 0.2 0.1 0 0.2 0.25 B. Murmann 0.3 0.35 L [μm] 0.4 0.45 0.5 EE214 Winter 2010-11 – Chapter 3 57 Extrinsic Capacitances (3) PMOS, gm/I D=10S/A, VDS=0.9V 0.8 Cggd/Cgg 07 0.7 Cdd/Cgg 0.6 05 0.5 0.4 0.3 0.2 0.1 0 B. Murmann 0.2 0.25 0.3 0.35 L [μm] 0.4 EE214 Winter 2010-11 – Chapter 3 0.45 0.5 58 Generic Design Flow 1) Determine gm (from design objectives) 2) Pick L Short channel Æ high fT (high speed) Long channel Æ high intrinsic gain 3) Pick gm/ID (or fT) Large gm/ID Æ low power, large signal swing (low VDSsat) Small gm/ID Æ high fT (high speed) 4) Determine ID (from gm and gm/ID) 5) Determine W (from ID/W, current density chart) Many other possibilities exist (depending on circuit specifics, design constraints and objectives) B. Murmann EE214 Winter 2010-11 – Chapter 3 59 Basic Design Example Given specifications and objectives – 0.18μm 0 18μm technology – DC gain = -4 – RL=1k, CL=50fF, Ri=10k – Maximize bandwidth while keeping IB ≤ 300uA • Implies L=Lmin=0.18um – Determine device width – Estimate dominant and nondominant pole B. Murmann EE214 Winter 2010-11 – Chapter 3 60 Small-Signal Model Calculate gm and gm/ID A v (0) ≅ gmRL = 4 B. Murmann ⇒ gm = 4 = 4mS 1kΩ EE214 Winter 2010-11 – Chapter 3 gm 4mS S = = 13.3 ID 300μA A 61 Why can we Neglect ro? A v (0) = gm (RL || ro ) ⎛ 1 1⎞ = gm ⎜ + ⎟ ⎝ RL ro ⎠ −1 1 1 1 = + A v (0) gmRL gmro 1 1 1 = + 4 gmRL gmro Even at L=Lmin= 0.18μm, we have gmro > 30 ro will be negligible in this design problem B. Murmann EE214 Winter 2010-11 – Chapter 3 62 Zero and Pole Expressions High frequency zero (negligible) ωz ≅ gm >> ωT Cgd Dominant pole (see Chapter 4) ωp1 ≅ 1 Ri ⎡⎣Cgs + Cgbb + (1 + gmRL ) ⋅ Cgdd ⎤⎦ Nondominant pole (see Chapter 4) ωpp2 ≅ 1 1 ωp1 Ri RL ⎡Cgs + Cgb ⎤ CL + ⎡Cgs + Cgd ⎤ Cdb + CLCgd ⎣ ⎦ ⎣ ⎦ ( Calculation of capacitances from tabulated parameters: Cgs + Cgb = Cgg − Cgd B. Murmann Cdb = Cdd − Cgd EE214 Winter 2010-11 – Chapter 3 63 Determine Cgg via fT Look-up L=0 18um L=0.18um 16.9 GHz B. Murmann EE214 Winter 2010-11 – Chapter 3 64 ) Find Capacitances and Plug in Cgg = Cgd = Cdd = 1 gm 1 4mS = = 37.7fF 37 7fF 2π fT 2π 16.9GHz Cgd Cgg Cgg = 0.24 ⋅ 37.7fF = 9.0fF Cdd Cgg = 0.60 ⋅ 39.4fF = 23.6fF Cgg ∴ fp2 ≅ 6.0 GHz ∴ fp11 ≅ 196 MHz B. Murmann EE214 Winter 2010-11 – Chapter 3 65 Device Sizing 16.2 A/m B. Murmann EE214 Winter 2010-11 – Chapter 3 L=0.18um 66 Circuit For Spice Verification Device width W= ID ID W = 300μA = 18.5μm 16.2A / m Simulation circuit o B i 1F B. Murmann EE214 Winter 2010-11 – Chapter 3 67 Simulated DC Operating Point element region id vgs vds vdsat vod gm gds ... cdtot cgtot cgd ... gm/ID B. Murmann 0:mn1 Saturati 326.8330u 624.9116m 873.1670m 113.7463m 138.5878m 138 5878m 4.1668m 108.2225u Calculation 300 uA 4 mS 21.8712f 37.6938f 8.9163f 23.6 fF 37.7 fF 9.0 fF 12.8 13.3 S/A EE214 Winter 2010-11 – Chapter 3 Good agreement! 68 HSpice .OP Capacitance Output Variables Corresponding Small Signal Model Elements HSpice (.OP) cdtot cgtot cstot cbtot cgs cgd B. Murmann ≡ Cgd + Cdb cgtot ≡ Cgs + Cgd + Cgb cstot ≡ Cgs + Csb bt t ≡ Cgb + Csb+ Cdb cbtot cgs ≡ Cgs cgd ≡ Cgd cdtot 21.8712f 37.6938f 44.2809f 34.9251f 26.7303f 8.9163f EE214 Winter 2010-11 – Chapter 3 69 Simulated AC Response 213 MHz 11.5 dB (3.8) 5.0 GHz Calculated values: |Av(0)|=12 dB (4.0), fp1=196 MHz, fp2=6.0 GHz B. Murmann EE214 Winter 2010-11 – Chapter 3 70 Using .pz Analysis Netlist statement .pz v(vo) vi Output *************************************************** ****** pole/zero analysis input = 0:vi output = v(vo) poles (rad/sec) real imag -1.34190g 0. -31.4253g 0. poles ( hertz) real imag -213.569x 0. -5.00149g 0. zeros (rad/sec) ( d/ ) real imag 458.247g 0. zeros ( h hertz) t ) real imag 72.9323g 0. B. Murmann EE214 Winter 2010-11 – Chapter 3 71 Observations The design and pole calculations essentially right on target! – Typical discrepancies are on the order of 10-20%, mostly due to VDS dependencies, finite output resistance, etc. We accomplished this by using pre-computed spice data in the design process Even if discrepancies are more significant, there’s always the possibility to track down the root causes – Hand calculations are based on parameters that also exist in Spice, e.g. gm/ID, fT, etc. – Different from square law calculations using μCox, VOV, etc. • Based on artificial parameters that do not exist or have no significance i ifi iin th the spice i model d l B. Murmann EE214 Winter 2010-11 – Chapter 3 72 References F. Silveira et al. "A gm/ID based methodology for the design of CMOS analog circuits and its application to the synthesis of a silicon-oninsulator micropower OTA, OTA," IEEE J. Solid Solid-State State Circuits, Sep. 1996, pp. 1314-1319. D. Foty, M. Bucher, D. Binkley, "Re-interpreting the MOS transistor via the inversion coefficient and the continuum of gms/Id," Proc. Int. Conf. on El t i Electronics, Circuits Ci it and dS Systems t , pp. 1179-1182, 1179 1182 S Sep. 2002 2002. B. E. Boser, "Analog Circuit Design with Submicron Transistors," IEEE SSCS Meeting, Santa Clara Valley, May 19, 2005, http://www ewh ieee org/r6/scv/ssc/May1905 htm http://www.ewh.ieee.org/r6/scv/ssc/May1905.htm P. Jespers, The gm/ID Methodology, a sizing tool for low-voltage analog CMOS Circuits, Springer, 2010. T. Konishi, K. Inazu, J.G. Lee, M. Natsu, S. Masui, and B. Murmann, “Optimization of High-Speed and Low-Power Operational Transconductance Amplifier Using gm/ID Lookup Table Methodology,” IEICE Trans. Electronics, Vol. E94-C, E94 C, No.3, Mar. 2011. B. Murmann EE214 Winter 2010-11 – Chapter 3 73 Chapter 4 Review of Elementary Circuit Configurations B. Murmann Stanford University y Reading Material: Sections 3.3.1, 3.3.3, 3.3.6, 3.3.8, 3.5, 7.2.3, 7.2.4.1, 7.3.2, 7.3.4, 4.2.2, 4.2.3, 4.2.4 Basic Single-Stage Amplifier Configurations MOS Common Source Common Gate Common Drain Common Emitter Common Base Common Collector Bipolar Transconductance Stage B. Murmann Current Buffer EE214 Winter 2010-11 – Chapter 4 Voltage Buffer 2 Widely Used Two-Transistor Circuits MOS Cascode Stage Current Mirror Differential Pair Bipolar p B. Murmann EE214 Winter 2010-11 – Chapter 4 3 Analysis Techniques (1) N Nodal d l analysis l i (KCL (KCL, KVL) – Write KCL for each node, solve for desired transfer function or port impedance – Most general method method, but conveys limited qualitative insight and often yields high-entropy expressions Miller theorem http://paginas.fe.up.pt/~fff/eBook/MDA/Teo_Miller.html Miller approximation pp − Approximate the gain across Z as frequency independent, i.e. K(s) ≅ K for the frequency range of interest − This approximation pp requires q a check ((or g good intuition)) B. Murmann EE214 Winter 2010-11 – Chapter 4 4 Analysis Techniques (2) Dominant D i t pole l approximation i ti 1 1 1 = ≅ 2 s s s s s2 ⎛ s ⎞⎛ s⎞ 1− + ⎜1 − ⎟⎜ 1 − ⎟ 1 − p − p + p p p1 p1p2 1 2 1 2 ⎝ p1 ⎠⎝ p2 ⎠ b 1 1 ⇒ p1 ≅ − , p2 ≅ − 1 Given 2 1 + b1s + b2s b1 b2 Zero value time constant analysis − The coefficient b1 can be found by summing all zero value time constants in the circuit b1 = ∑ τi Generalized time constant analysis − Can also find higher order terms (e.g. b2) using a sum of time constant products − A. Hajimiri, “Generalized Time- and Transfer-Constant Circuit Analysis,” IEEE Trans. Circuits Syst. I, pp. 1105-1121, June 2010. B. Murmann EE214 Winter 2010-11 – Chapter 4 5 Analysis Techniques (3) Return ratio analysis – See text pp. 599-612 Blackman’s impedance formula – See text pp. 607-612 Two Two-port port feedback analysis – See text pp. 557-587 – More later in this course B. Murmann EE214 Winter 2010-11 – Chapter 4 6 Chapter Overview BJT-centric review of elementary circuit configurations – Highlight differences to MOS circuits Single-stage amplifiers – Common emitter stage – Common emitter stage g with source degeneration g – Common collector stage – Common base stage – Common gate stage (discussion of bulk connection) BJT current mirrors BJT differential pair B. Murmann EE214 Winter 2010-11 – Chapter 4 7 Common-Emitter Stage VCC Vo RL RS vi ~ Q1 + VO+vo – IB VCC VI Vi Vi DC input bias voltage (VI) biases Q1 in the forward active region Typically, want VO ≅ VCC/2 Main differences to consider versus common-source stage (MOS) – Bias point sensitivity – Finite input resistance (due to rπ) – Base resistance (rb) often significant B. Murmann EE214 Winter 2010-11 – Chapter 4 8 Bias Point Sensitivity IB IC = βFIB RS VI + VBE – + R Vo L – Is e qVBE IB ≅ VI − VBE(on) VCC kT VO = VCC − ICRL = VCC − βFIBRL = VCC − βF RS RL ( VI − VBE(on) ) RS The dependence on βF makes “direct voltage biasing” impractical How to generate VI so as to control Vo? Practical configurations are usually based on feedback, replica biasing, ac coupling or differential circuits B. Murmann EE214 Winter 2010-11 – Chapter 4 9 Small-Signal Equivalent Circuit for CE Stage RS vi + v1 – ~ Cμ 1 rb rπ Cπ 2 gmv1 ro RL + CL vo – For hand analysis, we will usually neglect rc and re If significant, rb can be included with RS, i.e. RS* = RS + rb Resulting R lti llow-frequency f gain i A V (0) vo vi ω=0 ⎛ r ⎞ = − ⎜ * π ⎟⋅ gmRLtot ⎝ RS + rπ ⎠ RLtot Lt t = ro || RL =1 for MOS B. Murmann EE214 Winter 2010-11 – Chapter 4 10 Gate Tunnel Conductance for MOSFETS MOSFET MOSFETs with ith extremely t l thin thi gate t oxide id draw d a gate t currentt due d to t direct tunneling This leads to a finite current gain and input resistance – Similar to BJT! A Annema A. Annema, et al al., “Analog Analog circuits in ultra-deep-submicron ultra deep submicron CMOS CMOS,” IEEE J. J Solid Solid-State State Circuits, Circuits pp. pp 132-143 132 143, Jan. Jan 2005 2005. B. Murmann EE214 Winter 2010-11 – Chapter 4 11 Frequency Response Using nodal analysis, we find ⎛ s⎞ 1− ⎟ ⎜ ⎛ r ⎞ z1 ⎠ v o (s) = − ⎜ * π ⎟⋅ gmRLtot ⎝ v i (s) 1 + b1s + b2s2 ⎝ RS + rπ ⎠ b1 = RS* (Cπ + Cμ ) + RLtot (CL + Cμ ) + gmRS* RLtotCμ b2 = RS* RLtot (CπCL + CπCμ + CLCμ ) z1 = + gm Cμ z1 is a feedforward zero in the RHP. If Cπ >> Cμ, then z1 = + B. Murmann gm gm >> = ωT Cμ Cπ + Cμ EE214 Winter 2010-11 – Chapter 4 12 Dominant Pole Approximation If a dominant d i t pole l condition diti exists, i t we can write it p1 ≅ − 1 1 =− * b1 RS (Cπ + Cμ ) + RLtot (CL + Cμ ) + gmRS* RLtotCμ =− p2 ≅ − RS* 1 ⎡⎣Cπ + (1 + gmRLtot ) Cμ ⎦⎤ + RLtot (CL + Cμ ) RS* (Cπ + Cμ ) + RLtot (CL + Cμ ) + gmRS* RLtotCμ b1 =− b2 RS* RLtot ((CπCL + CπCμ + CLCμ ) If Cμ << Cπ, CL, then p2 ≅ − RS* Cπ + RLtotCL + gmRS* RLtotCμ RS* RLtotCπCL B. Murmann ⎛ 1 g C ⎞ 1 = −⎜ * + + m⋅ μ⎟ ⎝ RSCπ RLCL CL Cπ ⎠ EE214 Winter 2010-11 – Chapter 4 13 Emitter Degeneration VCC RL VO+vo RS vi VI Ro ~ Ri RE Gm ≅ gm 1 + gmRE Ro ≅ ro (1 + gmRE ) Degeneration resistor reduces the transconductance and increases the p resistance of the device output – Same as in the MOSFET version of this circuit For the BJT version, RE helps increase the input resistance B. Murmann EE214 Winter 2010-11 – Chapter 4 14 Calculation of Ri For the complete nodal analysis, see text p. 196 A more intuitive way to find Ri is via the Miller theorem K= Ri = ve RE g R ≅ = m E 1 vb + RE 1 + gmRE gm rπ rπ ≅ = rπ (1 + gmRE ) 1− K ⎛ gmRE ⎞ ⎜1− ⎟ ⎝ 1 + gmRE ⎠ The same “bootstrapping” effect applies to Cπ, we see Cπ/(1+gmRE) looking into the input − Assuming K = constant in the frequency range of interest B. Murmann EE214 Winter 2010-11 – Chapter 4 15 Alternative Calculation of Ri it it + vt - + v1 - ( +1)it RE + vt - RE( +1) + v1 - Ri ≅ rπ + RE (1 + β ) = rπ + RE (1 + gmrπ ) ≅ rπ (1 + gmRE ) Tricks of this kind are useful for reasoning g about low frequency q y behavior More detailed analyses must be used be taken when investigating frequency dependence B. Murmann EE214 Winter 2010-11 – Chapter 4 16 Small-Signal Equivalent Circuit for Degenerated CE Stage R S* vi Cμ + v1 – ~ rπ gmv1 Cπ + vE – R*S = RS + rb ro RL + vo – RE Deriving the transfer function of this circuit requires solving a 3x3 system of equations In order to obtain an estimate of the circuit’s bandwidth, it is more convenient (and intuitive) to perform a zero-value time constant analysis B. Murmann EE214 Winter 2010-11 – Chapter 4 17 Useful Expressions Ro ≅ ro R C + RE (f (for β → ∞ ) 1 + gmRE R π ≅ rπ RB + RE 1 + gmRE C B o E Rμ = Rleft + Rright + GmRleftRright Rleft ≅ RB rπ (1 + gmRE ) Rright ≅ RC Gm = B. Murmann EE214 Winter 2010-11 – Chapter 4 gm 1 + gmRE 18 Bandwidth Estimate for Degenerated CE Stage (1) R S* vi + v1 – ~ R*S Cμ rπ gmv1 Cπ + vE – = RS + rb RE ro RL + vo – (neglect) Rμ = RS* rπ (1 + gmRE ) + RC + Gm ⎡⎣RS* rπ (1 + gmRE ) ⎤⎦ RC ≅ RS* + RC + GmRCRS* = RS* (1 + A v (0) ) + RC R π ≅ rπ B. Murmann RS* + RE RS* + RE ≅ 1 + gmRE 1 + gmRE EE214 Winter 2010-11 – Chapter 4 19 Bandwidth Estimate for Degenerated CE Stage (2) RE RS* τ = ⎣⎡RS* (1 + A v (0) ) + RC ⎦⎤ Cμ + RS* Cπ 1 + gmRE 1+ ω−3dB ≅ 1 τ ω−3dB ≅ 1 τ Compare to the case of RE = 0 τ ≅ ⎡⎣RS* (1 + A v (0) ) + RC ⎤⎦ Cμ + RS* Cπ Adding RE can help improve the bandwidth, provided that gm > 1/RS* power dissipation p must be − Note,, however,, that gm ((and hence the p increased) to maintain the same Av(0) Consider another special case where gmRE>>1 and the time constant due to Cμ is negligible ⎛ R* ⎞ C τ ≅ ⎜1+ S ⎟ π ⎝ RE ⎠ gm B. Murmann ⎛ C + Cμ ⎞ ⎛ RE ⎞ ω−3dB ≅ ωT ⎜ π ⎟⎜ * ⎟ ⎝ C π ⎠ ⎝ RE + R S ⎠ EE214 Winter 2010-11 – Chapter 4 20 Common-Collector Stage (Emitter Follower) VCC RS vi Q1 ~ VO+vo VI IB RL CL Behavior is very similar to MOS common drain stage, except that – We do not need to worry about backgate effect – There is finite input resistance due to rπ – The output resistance depends on RS (in addition to 1/gm) B. Murmann EE214 Winter 2010-11 – Chapter 4 21 Input and Output Resistance Input I t resistance i t (by (b inspection) i ti ) Ri ≅ rπ (1 + gmRL ) Output O resistance i ((using i push-through h h h trick) i k) RS* = RS + rb Ro ≅ B. Murmann R* 1 1 ⎛ RS* ⎞ + S ≅ ⎜1+ ⎟ gm β + 1 gm ⎝ rπ ⎠ EE214 Winter 2010-11 – Chapter 4 22 Low Frequency Voltage Gain vi RS* vb vo Ri ≅ rπ (1 + gmRL ) A v0 = ≅ rπ (1 + gmRL ) vo vb vo gmRL = ≅ vi v i v b rπ (1 + gmRL ) + Rs* 1 + gmRL gmRL 1 + gmRL ≅1 B. Murmann RL for rπ (1 + gmRL ) >> Rs* for gmRL >> 1 and rπ (1 + gmRL ) >> R s* EE214 Winter 2010-11 – Chapter 4 23 Frequency Response Zi vo vb vo v Zi = ⋅ = ⋅ o vi v i v b Zi + R S v b Detailed analysis gives a very complex result for the general frequency response expression i Must typically apply approximations based on given component values B. Murmann EE214 Winter 2010-11 – Chapter 4 24 Frequency Response Assuming that RS is large (often the case, and the reason why the stage is used), we expect that the dominant pole is introduced at node vb For the frequency q y range g up p until the dominant p pole,, we can therefore approximate vo gmRL ≅ =K v b 1 + gmRL Zi ≅ ωp ≅ 1 1 = s ( Cπ [1 − K ] + Cμ ) sCi 1 (Rs Rin ) Ci See text, pp. 503 for a more detailed analysis, which also captures the feedforward zero introduced by Cπ B. Murmann EE214 Winter 2010-11 – Chapter 4 25 Common Collector Output Impedance Detailed analysis of the common-collector output impedance shows potentially inductive behavior for large RS The inductive behavior can lead to undesired “ringing” ringing (or oscillations) during circuit transients In other cases, the inductive behavior is utilized for bandwidth extension ((“inductive inductive peaking”) peaking ) The capacitance Cμ tends to reduce the inductive frequency range – Cμ appears in parallel with RS, and creates a low impedance termination for high frequencies – Makes it difficult to use the circuit as a “good inductor” For a discussion on common collector output impedance and a detailed KCL-based analysis, see EE114 or section 7.2.3 B. Murmann EE214 Winter 2010-11 – Chapter 4 26 Common-Base Stage VCC Ai = RL io Ri Ii + ii Ro VO+vo io ii Ai (0) = iC β = iE β + 1 Neglecting rb, rc, re and rπ, we have Ro ≅ ro (1 + gmRS ) RS (large) Ri ≅ 1 ⎛ RL ⎜1 + gm ⎝ ro ⎞ 1 ⎟≅ ⎠ gm Behavior is very similar to MOS common base stage stage, except that – We do not need to worry about backgate effect – The DC current gain is not exactly unity, due to finite β B. Murmann EE214 Winter 2010-11 – Chapter 4 27 Simplified Small-Signal Model for High-Frequency Analysis vo RL = ii ⎛ s ⎞⎛ s ⎞ ⎜ 1 − ⎟⎜ 1 − ⎟ ⎝ p1 ⎠⎝ p2 ⎠ ωp2 = gm Cπ + Cμ = ωT Cπ Cπ ωp1 = 1 RL CL The time constant associated with the load usually dominates the frequency response, i.e. ωp1 < ωp2 Note, however, that ωp2 can be important in feedback circuits (phase margin) B. Murmann EE214 Winter 2010-11 – Chapter 4 28 Common Gate Stage – Bulk Connection Scenarios Cgb Cgb Cdb Cdb Cbsub (DWELL) Cbsub (DWELL) VDD Csb Cgs VDD ii ωp2,a = Csb Cgs ii gm Cggs + Cggb + Cbsub + Cdb [1 − K(s)] ωp2,b = gm + gmb Cggs + Csb ωp2,a is always less than ωp2,b Æ Usually a bad idea to connect source to gate stage g bulk in a common g B. Murmann EE214 Winter 2010-11 – Chapter 4 29 Backgate Effect in the EE214 Technology (1) B. Murmann EE214 Winter 2010-11 – Chapter 4 30 Backgate Effect in the EE214 Technology (2) B. Murmann EE214 Winter 2010-11 – Chapter 4 31 Basic BJT Current Mirror IOUT = IC2 IIN Q1 + VBE – Q2 + Vo = VCE2 – IC2 = IC1 IS2e IS1e VBE VT VBE VT ⎛ VCE2 ⎞ ⎜1+ ⎟ VA ⎠ IS2 ⎛ VCE2 VCE1 ⎞ ⎝ ≅ − ⎜1+ ⎟ I V VA ⎠ ⎝ S1 A ⎛ VCE1 ⎞ ⎜1+ ⎟ VA ⎠ ⎝ Error due to base current IIN = IC1 + IB1 + IB2 = IC1 + IC1 IC2 I ⎞ ⎛ + ≅ IC2 ⎜ 1 + 2 C2 ⎟ β β β ⎠ ⎝ for IS1 = IS2 IOUT 1 2 ≅ ≅ 1− IIN β ⎛ 2⎞ ⎜1+ β ⎟ ⎝ ⎠ B. Murmann EE214 Winter 2010-11 – Chapter 4 32 BJT Current Mirror with “Beta Helper” VCC IIN IOUT Q3 + IE3 Q1 − IE3 = Q2 VOUT IC1 IC2 I + ≅ 2 C2 β β β assumin g IS1 = IS2 – IB3 = − IE3 2IC2 ≅ β + 1 β(β + 1) IIN = IC1 + IB3 ≅ IC1 + IOUT ≅ IIN B. Murmann 1 ⎛ 2 1+ ⎜ 2 ⎜β +β ⎝ ⎞ ⎟⎟ ⎠ ≅ 1− 2IC2 ⎡ 2 ⎤ ≅ IC2 ⎢1 + ⎥ β(β + 1) ⎣ β(β + 1) ⎦ 2 β2 EE214 Winter 2010-11 – Chapter 4 33 BJT Current Mirror with Degeneration IIN IOUT + Q1 Q2 R1 IC2 = R2 VOUT Neglecting base currents VBE1 + IC1R1 = VBE2 + IC2R2 – ⎡⎛ IC1 ⎞ ⎛ IS2 ⎞ ⎤ ⎫⎪ R1 1 ⎧⎪ ⎨IC1R1 + VT ln ⎢⎜ ⎟ ⎜ ⎟ ⎥⎬ ≅ IC1 R2 ⎩⎪ R2 ⎣⎝ IC2 ⎠ ⎝ IS1 ⎠ ⎦ ⎭⎪ IOUT R1 ≅ IIN R2 Degeneration brings two benefits – Increased output resistance – Reduces sensitivity of mirror ratio to mismatches in IS Minimum VOUT for which Q2 remains forward active is increased B. Murmann EE214 Winter 2010-11 – Chapter 4 34 Differential Circuits Vi1 + – Vo1 Vi2 – + Vo2 Enables straightforward biasing without AC coupling Information is carried in “differential” signals that are insensitive to “common-mode” perturbations, such as power supply noise Fully differential circuits are increasingly used for I/O and clock-and-data recovery (CDR) circuits to allow the use of small signals at high speeds B. Murmann EE214 Winter 2010-11 – Chapter 4 35 BJT Differential Pair VCC RC1 Vo1 Vi1 Differential Input Voltage RC2 IC1 Q1 Vid Vi1 − Vi2 Vo2 IC2 Q2 Vi2 Differential Collector Current Icd Ic1− Ic2 Differential Output Voltage ITAIL RTAIL Vod Vo1 − Vo2 VEE The following large signal analysis neglects rb, rc, re, finite REE and assumes that the circuit is perfectly symmetric B. Murmann EE214 Winter 2010-11 – Chapter 4 36 Large Signal Analysis IC1 ≅ IS1 e Vi1 − Vbe1 + Vbe2 − Vi2 = 0 I ⇒ c1 = e Ic2 ITAIL = − (Ie1+ Ie2) = Vbe1 − Vbe2 VT 1 (Ic1+ Ic2) α =e Vbe1 VT Vi1 − Vi2 VT IC2 ≅ IS2 e =e Vid VT αITAIL ⇒ Ic1 = 1+ e ⎡ ⎢ 1 1 Icd = Ic1 − Ic2 = αFITAIL ⎢ − Vid V − + id ⎢ VT 1 + e VT ⎣1 + e Vbe2 VT V − id VT Ic2 = αITAIL 1+ e V + id VT ⎤ ⎛ Vid ⎞ ⎥ ⎥ = αFITAIL tanh ⎜ 2V ⎟ ⎝ T⎠ ⎥ ⎦ ⎛ V ⎞ Vod = IodRL = αITAILRL tanh ⎜ id ⎟ ⎝ 2VT ⎠ B. Murmann EE214 Winter 2010-11 – Chapter 4 37 Plot of Transfer Characteristics Text, p. 215 Text, p. 216 Linear region in Vod vs. vs Vid characteristic is narrow compared to MOS – Recall that full steering in a MOS pair occurs for Vid = 2VOV BJT differential pair is linear only for |Vid| < VT ≅ 26 mV B. Murmann EE214 Winter 2010-11 – Chapter 4 38 Emitter Degeneration C Can use emitter itt degeneration d ti resistors i t tto iincrease th the range off iinputt voltage over which the transfer characteristic of the pair is linear For large RE, linear range is approximately equal to ITAILRE VCC RC RC Text, p. 217 Vo2 Vo1 Q1 Vi1 Q2 RE Vi2 RE ITAIL VEE B. Murmann EE214 Winter 2010-11 – Chapter 4 39 Voltage Decomposition Common-Mode Voltages VCC 1 Vic (Vi1 + Vi2 ) 2 1 Voc (Vo1 + Vo2 ) 2 RC1 Voc+Vod/2 Inputs Vi1 and Vi2 can be decomposed into a combination of differentialdifferential and common common-mode mode voltage sources 1 Vi1 = Vic + Vid 2 1 Vi2 = Vic − Vid 2 B. Murmann RC2 Voc–Vod/2 Q1 Vid/2 Q2 –V Vid/2 ITAIL VEE Vic EE214 Winter 2010-11 – Chapter 4 40 Small Signal Model for vic = 0 Text, p. 224 Define differential mode gain as B. Murmann A dm v od v id vic = 0 EE214 Winter 2010-11 – Chapter 4 41 Differential Mode Half Circuit Text, p. 225 v od v = −gmR id 2 2 B. Murmann A dm = v od = −gmR v id EE214 Winter 2010-11 – Chapter 4 42 Small Signal Model for vid = 0 Text, p. 227 Define common mode gain as B. Murmann A cm v od v id vid = 0 EE214 Winter 2010-11 – Chapter 4 43 Common Mode Half Circuit Text, p. 227 v oc = −GmRv ic B. Murmann A cm = v oc gmR = −GmR = − v ic 1 + 2gmRTAIL EE214 Winter 2010-11 – Chapter 4 44 Interaction of Common Mode and Differential Mode A cdm v od v ic i A dcm and vid = 0 v oc v id vic = 0 In a perfectly balanced (symmetric) circuit, Acdm = Adcm = 0 In practice, Acdm and Adcm are not zero because of component mismatch Acdm is i iimportant t t because b it iindicates di t th the extent t t tto which hi h a commonmode input will corrupt the differential output (which contains the actual signal information) See S ttext, t section ti 3.5.6.9 3 5 6 9 ffor a d detailed t il d analysis l i B. Murmann EE214 Winter 2010-11 – Chapter 4 45 Common-Mode Rejecton For fully differential circuits, the common-mode rejection ratio (CMRR) is traditionally defined as CMRR A dm A cdm However, the text defines the ratio as CMRR Text T A dm A cm This latter definition is appropriate for circuits with a differential input and single-ended output, such as operational amplifiers. B. Murmann EE214 Winter 2010-11 – Chapter 4 46 Input-Referred DC Offsets In I a perfectly f tl symmetric t i circuit, i it Vid = 0 yields i ld Vod = 0 Imbalances can be modeled as input referred offsets VCC VCC RC1 Vo2 Vo1 IB1 Vi1 RC RC2 Q1 Vo2 Vo1 Q2 – Vi1 Vi2 Vos + Q1 Q2 Ios/2 Vi2 IB2 RC IEE IEE OFFSETS VEE PAIR W/ MISMATCH B. Murmann VEE IDEAL PAIR EE214 Winter 2010-11 – Chapter 4 47 Analysis Vos − VBE1 + VBE2 = 0 ⎛I ⎞ ⎛I ⎞ ∴ Vos = VT ln ⎜ C1 ⎟ − VT ln ⎜ C2 ⎟ ⎝ IS1 ⎠ ⎝ IS2 ⎠ ⎡⎛ I ⎞ ⎛ I ⎞ ⎤ = VT ln ⎢⎜ C1 ⎟ ⎜ S2 ⎟ ⎥ , ⎢⎣⎝ IC2 ⎠ ⎝ IS1 ⎠ ⎥⎦ where VT = kT q If Vod = 0, then IC1RC1 = IC2RC2 ∴ Thus B. Murmann IC1 IC2 = RC2 RC1 ⎡⎛ R ⎞ ⎛ I ⎞ ⎤ Vos = VT ln ⎢⎜ C2 ⎟ ⎜ S2 ⎟ ⎥ ⎢⎣⎝ RC1 ⎠ ⎝ IS1 ⎠ ⎥⎦ EE214 Winter 2010-11 – Chapter 4 48 Result For small mismatches ΔRC << RC and ΔIS << IS, it follows that −1 Vos ⎛ ΔRC ΔIS ⎞ ⎛ gm ⎞ ⎛ ΔRC ΔIS ⎞ ≅ VT ⎜ − − − ⎟=⎜ ⎟ ⎟ ⎜− IS ⎠ ⎝ ID ⎠ ⎝ RC IS ⎠ ⎝ RC And similarly Ios ≅ − IC ⎛ ΔRC Δβ ⎞ + ⎜ ⎟ β ⎝ RC β ⎠ (see text, pp. 231) Mismatch in IS results primarily from mismatches in the emitter areas and the base doping Mismatch in β results primarily from mismatches in the base width The standard deviations of device-to-device variations in IS and β is typically on the order of 5% B. Murmann EE214 Winter 2010-11 – Chapter 4 49 Offset Voltage Drift dVos d ⎡ kT ⎛ ΔRC ΔIS ⎞ ⎤ Vos = − − ⎢ ⎥= dT IS ⎟⎠ ⎥⎦ T dT ⎢⎣ q ⎜⎝ RC Example – VOS was determined d t i d tto b be 2 mV V th through h a measurementt att 300°K – Means that the offset voltage will drift by 2 mV/300°K = 6.6 μV/°C For a MOS differential pair, the offset drift is less predictable and turns outt to t be b a complex l function f ti off severall process parameters t B. Murmann EE214 Winter 2010-11 – Chapter 4 50 Comparison of VOS for MOS and BJT Differential Pairs −1 VOS,BJT VOS,MOS ⎛ g ⎞ ⎛ ΔR ΔIS ⎞ ≅ ⎜ m ⎟ ⎜− − ⎟ IS ⎠ ⎝ ID ⎠ ⎝ R ⎛g ⎞ ≅ ΔVt + ⎜ m ⎟ ⎝ ID ⎠ ? −1 ⎛ ΔR Δ ( W / L ) ⎞ − ⎜⎜ − ⎟ R ( W / L ) ⎟⎠ ⎝ Worse Similar to BJT The standard deviation of ΔVt can be estimated using the following expression (Pelgrom, JSSC 10/1989) σΔVt ≅ A Vt WL where Avt ≅ 5mV-μm for a typical 0.18 μm process B. Murmann EE214 Winter 2010-11 – Chapter 4 51 Numerical Example Ignoring I i resistor i t mismatch i t h ffor simplicity i li it Assume (gm/ID)MOS = 10 S/A, W = 5 μm, L= 0.2 μm ⎡⎛ g ⎞−1 ⎛ ΔI ⎞ ⎤ std ( VOS,BJT ) ≅ std ⎢⎜ m ⎟ ⎜ S ⎟ ⎥ = 26mV ⋅ 5% = 1.3mV ⎢⎝ ID ⎠ ⎝ IS ⎠ ⎥ ⎣ ⎦ −1 ⎡ ⎛ gm ⎞ ⎛ Δ ( W / L ) ⎞ ⎤ ⎢ std ( VOS,MOS ) ≅ std ΔVt + ⎜ ⎟⎟ ⎥ ⎟ ⎜⎜ I W / L ( ) ⎢ ⎝ D ⎠ ⎝ ⎠ ⎥⎦ ⎣ 2 ⎛ 5mV − μm ⎞ 2 ≅ ⎜ + 100mV ⋅ 5% ) ⎜ 5μm ⋅ 0.2μm ⎟⎟ ( ⎝ ⎠ ≅ ( 5mV )2 + ( 5mV )2 = 7.1mV MOS offset is typically 5 5-10 10 times worse than BJT B. Murmann EE214 Winter 2010-11 – Chapter 4 52 AVt Data AVt improves with technology scaling [Chang, TED 7/2005] But, unfortunately, AVt scales not as fast as minimum device area – Hence H Vt mismatch i t h ffor minimum i i size i d devices i worsens B. Murmann EE214 Winter 2010-11 – Chapter 4 53 Ch t 5 Chapter Two-Port Two Port Feedback Circuit Analysis B Murmann B. M Stanford University Reading Material: Sections 8.1, 8.2, 8.3, 8.4, 8.5, 8.6, 8.8 Benefits and Costs of Negative Feeback Negative feedback provides a means of exchanging gain for improvements in other performance metrics Benefits • Reduced sensitivity (improved precision) • Reduced distortion (more later) • Scaling of impedance levels (up or down) • Increased bandwidth Costs B. Murmann • Lower gain • Potential instability EE214 Winter 2010-11 – Chapter 5 2 Ideal Feedback (1) Si + Σ Sε a – So Sfb f Assumptions for an ideal feedback system: 1. No loading 2. Unilateral transmission in both the forward amplifier and feedback network So = a ⋅ Sε Sfb = f ⋅ So ⇒ So = (Si − Sfb ) = a(Si − f ⋅ So ) Sε = Si − Sfb B. Murmann EE214 Winter 2010-11 – Chapter 5 3 Ideal Feedback (2) Closed-Loop Gain: A Loop Gain: T So a = Si 1 + af af = Sfb Sε ⇒ A= a 1+ T If T >> 1, then A≅ a 1 = T f The feedback loop acts to minimize the error signal, Sε, thus forcing Sfb to t track t k Si. In I particular, ti l ⎛ a ⎞ ⎛ af ⎞ Sε = Si − f ⋅ So = Si − f ⋅ ⎜ Si = ⎜ 1 − ⋅S ⎟ ⎝ 1 + aff ⎠ ⎝ 1 + aff ⎟⎠ i ∴ B. Murmann Sε Si = 1− T 1 = 1+ T 1+ T and ⎛S ⎞ Sfb T = a⋅f⎜ ε ⎟ = Si ⎝ Si ⎠ 1+ T EE214 Winter 2010-11 – Chapter 5 4 Gain Sensitivity Th The ffeedback db k network t k is i ttypically i ll a precision i i passive i network t k with ith an insensitive, well-defined transfer function f. The forward amplifier gain is generally large, but not well controlled. F Feedback db k acts t to t reduce d nott only l the th gain, i b butt also l th the relative, l ti or fractional, gain error by the factor 1+T dA d ⎛ a ⎞ 1 d ⎛ 1 ⎞ = +a ⎜ = ⎜ ⎟ da da ⎝ 1+ af ⎠ 1+ af da ⎝ 1+ af ⎟⎠ = (1+ af) − af (1+ af) 2 = 1 (1+ af) 2 = 1 (1+ T)2 For a change δa in a δ = δa ∴ B. Murmann dA δa δ = δa da (1+ T)2 δA δa ⎛ 1+ T ⎞ ⎛ 1 ⎞ δa = = A (1+ T)2 ⎜⎝ a ⎟⎠ ⎜⎝ 1+ T ⎟⎠ a EE214 Winter 2010-11 – Chapter 5 5 The Two-Port Approach to Feedback Amplifier Design A practical approach to feedback amplifier analysis and design is based on constructing two-port representations of the forward amplifier and feedback network The two two-port port approach relies on the following assumptions: – Loading effects can be incorporated in the two-port model for the forward amplifier – Transmission through the forward amplifier is nearly unilateral – Forward transmission through the feedback network is much less than that through the forward amplifier When the preceding assumptions break down, the two-port approach deviates from from, and is less accurate than than, the return ratio approach devised by Henrik Bode But, the two-port approach can provide better physical tuition, especially with respect p to what happens pp in the frequency q y domain when the feedback loop is “closed” – It is basically an effort to model feedback amplifiers in the way that Harold Black conceived them B. Murmann EE214 Winter 2010-11 – Chapter 5 6 Feedback Configurations IIn the th two-port t t approach h to t feedback f db k amplifier lifi analysis l i th there are ffour possible amplifier configurations, depending on whether the two-port networks are connected in SHUNT or in SERIES at the input and output p of the overall amplifier At the OUTPUT – A shunt connection senses the output voltage – A series connection senses the output current At the INPUT – A shunt connection feeds back a current in parallel with the input – A series connection feeds back a voltage in series with the input The four possible configurations are referred to as SERIES-SHUNT, SERIES SHUNT, SHUNT-SHUNT, SHUNT-SERIES, and SERIES-SERIES feedback The following pages illustrate these configurations using ideal two-port networks B. Murmann EE214 Winter 2010-11 – Chapter 5 7 Series-Shunt Feedback + + a vε – + vo – vi – + + – – vfb f a= vo v , f = fb vε vo A= vo a a = = v i 1+ af 1+ T In this case a, A and f are all voltage gains B. Murmann EE214 Winter 2010-11 – Chapter 5 8 Shunt-Shunt Feedback iε ii + vo a – ifb f a= vo i , f = fb iε vo A= vo a a = = 1+ af 1+ T ii a and A are transimpedances; f is a transadmittance B. Murmann EE214 Winter 2010-11 – Chapter 5 9 Shunt-Series Feedback iε ii a io ifb f a= io i , f = fb iε io A= io a a = = ii 1+ af 1+ T a, A, and f are current gains B. Murmann EE214 Winter 2010-11 – Chapter 5 10 Series-Series Feedback + + a vε io – vi – + + – – vfb f a= io v , f = fb vε io A= io a a = = v i 1+ af 1+ T a and A are transadmittances; f is transimpedance Note: T = af is always y dimensionless B. Murmann EE214 Winter 2010-11 – Chapter 5 11 Closed Loop Impedances T To illustrate ill t t the th influence i fl off feedback f db k on the th input i t and d output t t impedances i d off an amplifier, consider the following: – Include finite input and output impedances in a simple, idealized two-port model of the forward amplifier – Assume that the feedback network has ideal input and output impedances so as not to load the forward amplifier Consider two examples, series-shunt and shunt-series amplifiers SERIES-SHUNT io vε B. Murmann avε EE214 Winter 2010-11 – Chapter 5 12 “Ideal” Series-Shunt Impedances Input Impedance vi ii Zi O t t Impedance Output I d Zo io = 0 With io = 0 vi = 0 With vi = 0 v o = av ε v ε + fv o = v i = 0 v i = v ε + fv o = (1+ af)v ε io = = (1+ T)v ε ii = vo io vε zi Zi = = 1⎛ 1 ⎞ v zi ⎜⎝ 1+ T ⎟⎠ i = vi = (1+ T)z i ii v o − av ε zo 1 1+ af v o zo ( ) 1 (1+ T)v o zo Zo = B. Murmann = vo z = o io 1+ T EE214 Winter 2010-11 – Chapter 5 13 “Ideal” Shunt-Series Impedances Basic amplifier p iε zi ii io + aiε zo vo ifb f io Input Impedance Zi vi ii With vo = 0 – v o =0 io = aiε i i= iε + fio = (1+ af) iε = (1+ T)iε ⇒ ⎛ i ⎞ v i = iε zi = ⎜ i ⎟ zi ⎝ 1+ T ⎠ B. Murmann EE214 Winter 2010-11 – Chapter 5 Zi = vi zi = i i (1+ T) 14 O t t Impedance Output I d Zo With ii = 0 vo io i i= 0 iε + fio = 0 v o = (io + aiε )zo = (io + afio )zo ⇒ Zo = = io (1+ T)zo vo = (1+ T)zo io In general: • Negative feedback connected in series increases the driving point impedance by (1+T) • Negative feedback connected in shunt reduces the driving point impedance by (1+T) B. Murmann EE214 Winter 2010-11 – Chapter 5 15 Loading Effects – Introductory Example Consider C id th the ffollowing ll i ffeedback db k circuit i it Analysis methods − Closed loop transfer function using nodal analysis − Return ratio analysis (see EE114) − Two-port feedback circuit analysis B. Murmann EE214 Winter 2010-11 – Chapter 5 16 Nodal Analysis Given v1 − vo 0 − ii RF vo − v1 0 + g m⋅ v 1 + RF vo RL ⎛⎜ RF⋅ ii + RL⋅ ii ⎞⎟ RL⋅ g m + 1 ⎜ ⎟ Find( v 1 , v o ) → ⎜ ⎟ ⎜ RL⋅ ii − RF⋅ RL⋅ g m⋅ ii ⎟ ⎜ ⎟ RL⋅ g m + 1 ⎝ ⎠ A vo RL − RF⋅ RL⋅ g m ii RL⋅ g m + 1 ⎛1− ⎜ −RF⋅ ⎜ ⎜ ⎜1+ ⎝ ⎞ ⎟ ⎟ 1 ⎟ g m⋅ RL ⎟ ⎠ 1 g m⋅ RF No information about loop gain (which we need e.g. for stability analysis) B. Murmann EE214 Winter 2010-11 – Chapter 5 17 Return Ratio Analysis A A A inf ⋅ −RF RR 1 + RR + g m⋅ RL 1 + g m⋅ RL d A inf 1 + RR + RL 1 + g m⋅ RL −RF d RL R ⎞ ⎛ ⎛1− ⎜ g m⋅ RL − L ⎟ ⎜ R ⎜ F⎟ −RF⎜ −RF⋅ ⎜ ⎜ 1 + g m⋅ RL ⎟ ⎝ ⎠ ⎜1+ ⎝ RR g m⋅ RL ⎞ ⎟ ⎟ 1 ⎟ g m⋅ RL ⎟ ⎠ 1 g m⋅ RF The result for the closed loop gain (A) matches the nodal analysis perfectly In addition, addition we have determined (along the way) the loop gain (RR) – This is useful for stability analysis The return ratio method is accurate and general The two-port method aims to sacrifice some of this accuracy and generality in exchange for better intuition and less computational effort – The involved approximations follow from the typical design intent for each of the four possible approximations B. Murmann EE214 Winter 2010-11 – Chapter 5 18 Shunt-Shunt Feedback Model iε ii + vo a – ifb f RF feeds back a current that is proportional to the output voltage – The appropriate two-port model to use for this amplifier is therefore the “shunt-shunt” configuration We wish to identify proper “a” a and “ff “ blocks that model our circuit The key issue is that “a” and “f” are not perfectly separable without making any approximations – RF is responsible for feedback feedback, but also affects the behavior of “a” a B. Murmann EE214 Winter 2010-11 – Chapter 5 19 y-Parameter Model for the Feedback Network (1) B. Murmann EE214 Winter 2010-11 – Chapter 5 20 y-Parameter Model for the Feedback Network (2) y11f = 1 RF y 21f = − B. Murmann 1 RF y12f = − 1 RF y 22f = 1 RF EE214 Winter 2010-11 – Chapter 5 21 y-Parameter Model for the Feedback Network (3) Final steps – Absorb y11 and y22 into forward amplifier – Neglect feedforward through feedback network (y21) • This is justified by the usual design intent: we do not want the feedforward through the feedback network to be significant! B. Murmann EE214 Winter 2010-11 – Chapter 5 22 Identification of “a” and “f” iε ii + vo a – ifb f f A B. Murmann − 1 a RF 1 f ⋅ 1 1+ 1 aff vo v1 ⋅ v 1 ie −g m⋅ RL⋅ RF RL + RF ⋅ RF af g m⋅ RL⋅ RF RL + RF 1 RF 1+ 1 RF + RL ⋅ g m RF⋅ RL EE214 Winter 2010-11 – Chapter 5 23 Comparison and Generalization Note that for RL << RF (i.e. small feedforward) – The product of a and f approaches the true loop gain (RR) – The two-port closed-loop gain expression (A) approaches the nodal analysis and return ratio result The approach illustrated in the previous example can be generalized to cover all four feedback configurations For all four configurations, configurations the following analysis steps apply – Identify input and output variables and feedback configuration – Find feedback function “f” – Add loading impedances to input and output port of the basic amplifier “a” – Perform calculations using ideal two-port feedback equations B. Murmann EE214 Winter 2010-11 – Chapter 5 24 B. Murmann EE214 Winter 2010-11 – Chapter 5 25 General Analysis for Shunt-Shunt Feedback Forward amplifier p iS B. Murmann EE214 Winter 2010-11 – Chapter 5 26 Summing currents at the input and output of the overall amplifier iS = (yS + y11a + y11f )v i + (y12a + y12f )v o 0 = (y21a + y21f )v i + (yL + y22a + y22f )v o Define yi y S + y11a + y11f yo yL + y 22a + y 22f Then vi = − yo v y21a + y21f o ⎛ ⎞ −yo iS = yi ⎜ ⎟ v o + (y12a + y12f )v o ⎝ y21a + y21f ⎠ ⎛ ⎞ 1 ⎡⎣ −yiyo + (y21a + y21f )(y12a + y12f ) ⎤⎦ v o =⎜ ⎟ ⎝ y21a + y21f ⎠ B. Murmann EE214 Winter 2010-11 – Chapter 5 ∴ 27 −(y vo (y21a + y21f ) = iS yiyo − (y21a + y21f )(y12a + y12f ) ⎛ y + y21f ⎞ − ⎜ 21a ⎟ yiyo ⎝ ⎠ = ⎛ y + y21f ⎞ 1− ⎜ 21a ⎟ (y12a + y12f ) yiyo ⎝ ⎠ Comparing this result with the ideal feedback equation vo a =A= is 1+ af it is apparent that a feedback representation can be used by defining B. Murmann y 21a + y 21f yi y o a − f y12a + y12f EE214 Winter 2010-11 – Chapter 5 28 For the preceding definitions of a and ff, it is difficult to establish equivalent circuits for these functions. However, the situation can be simplified by realizing that it may often be possible to neglect reverse transmission in the forward amplifier and forward transmission in the feedback network. That is, it can often be assumed that y12a << y12f and y21a >> y21f in which case ⎛ −y21a ⎞ ⎜ yy ⎟ vo ⎝ i o ⎠ A= = iS ⎛ −y ⎞ 1+ ⎜ 21a ⎟ y12f ⎝ yiyo ⎠ B. Murmann ⇒ a=− y21a yiyo f = y12f EE214 Winter 2010-11 – Chapter 5 29 For the simplified definitions, definitions it is possible to identif identify separate eq equivalent i alent circuits for a and f : yi New forward amplifier yo iS yi = yS + y11a + y11f yo = yL + y22a + y22f New feedback network The admittances yS, y11a, y22f and yL have been “pulled” into the forward amplifier. Basically, the “loading” of the feedback network, as well as the source and load admittances, is absorbed in the equivalent forward amplifier. B. Murmann EE214 Winter 2010-11 – Chapter 5 30 Example – Op Amp w/ Shunt-Shunt Feeback iS Small-signal equivalent circuit: Feedback network ro iS B. Murmann ri EE214 Winter 2010-11 – Chapter 5 31 I order In d to d determine i the h lloading di off the h ffeedback db k network k on the h fforward d amplifier, the y parameters of the feedback network are first determined using the following circuit : y11f = y22f = y12f = i1 v1 i2 v2 i1 v2 = 1 RF = 1 RF v 2 =0 v1 =0 =− v1 =0 1 =f RF Also, y21f = –1/RF, but it is neglected in comparison with y21a B. Murmann EE214 Winter 2010-11 – Chapter 5 32 Th overallll circuit The i i can now b be separated d iinto fforward d and d ffeedback db k paths, h with the loading of the feedback network included within the forward path Forward amplifier f ro iS ri B. Murmann EE214 Winter 2010-11 – Chapter 5 33 To determine the forward amplifier gain gain, a a, break the feedback loop at the “output” of the feedback network by setting ifb = 0. Then ⎛ r i RF ⎞ v1 = (r i PRF )iS = ⎜ ⎟ iS ⎝ r i + RF ⎠ ⎛ R ⎞ vo = − ⎜ ⎟ a v v1 ⎝ R + ro ⎠ Thus a= vo iS i fb = 0 where R RF RL ⎛ r i RF ⎞ ⎛ R ⎞ = −a av ⎜ ⎟⎜ ⎟ ⎝ r i + RF ⎠ ⎝ R + ro ⎠ Since f = –1/RF, the loop p gain, g , T = af,, is ⎛ r i ⎞ ⎛ R PR ⎞ ⎛ ri ⎞ ⎛ ⎞ RLRF F L T = av ⎜ ⎟⎜ ⎟ = av ⎜ ⎟⎜ ⎟ ⎝ r i + RF ⎠ ⎝ RF PRL + ro ⎠ ⎝ r i + RF ⎠ ⎝ RLRF + roRF + roRL ⎠ B. Murmann EE214 Winter 2010-11 – Chapter 5 34 The input and output impedances of the equivalent forward amplifier are zia = RF || ri zoa = ro || RF || RL Since the feedback network is connected in shunt at both the input and output of the forward amplifier, the closed-loop input and output impedances are RF || ri zia i = 1+ T 1+ T zoa ro || RF || RL Zo = = 1+ T 1+ T Zi = B. Murmann EE214 Winter 2010-11 – Chapter 5 35 Series-Series Feedback In a series-series feedback amplifier, p the forward amplifier p and feedback network share common currents at the input and output. Therefore, open circuit impedance parameters (z parameters) are used to characterize the two-port networks. v1 = z11 i1 + z12 i2 v 2 = z 21 i1 + z 22 i2 where z11 v1 i1 i z12 v2 i1 z22 2 =0 z21 B. Murmann i2 = 0 v1 i2 i 1=0 v2 i2 EE214 Winter 2010-11 – Chapter 5 i1 = 0 36 Series-Series Feedback, cont’d The two-port representation of a series-series feedback amplifier is then Forward amplifier io ii vS B. Murmann EE214 Winter 2010-11 – Chapter 5 37 Series-Series Feedback, cont’d Summing voltages at the input and output v S = (z S + z11a + z11f )ii + (z12a + z12f )io 0 = (z ( 21a + z 21f )i i + (z ( L + z 22a + z 22f )io Define zi zS + z11a + z11f zo zL + z22a + z22f Again, g , neglect g reverse transmission through g the forward amplifier p and feedforward through the feedback network; that is, assume z12a << z12f and z 21a >> z 21f B. Murmann EE214 Winter 2010-11 – Chapter 5 38 Series-Series Feedback Equations Under the preceding assumptions, the series-series feedback amplifier can be represented by a two-port model with ⎛ −z ⎞ 21a ⎜ ⎟ io ⎝ z i zo ⎠ a A= = = vS ⎛ −z 1+ af z ⎞ 1+ ⎜ 21a ⎟ z12f ⎝ z i zo ⎠ where a=− z 21a z i zo f = z12f B. Murmann EE214 Winter 2010-11 – Chapter 5 39 Series-Series Feedback Equivalent Two-Port Networks Equivalent two two-port port networks can now be identified for a and f in which the loading of the feedback network is included in the forward amplifier New forward amplifier vS B. Murmann EE214 Winter 2010-11 – Chapter 5 40 Series-Series Triple io AC circuit : Q3 Q2 RS + vS ~ – Q1 ii RC1 iE1 RC2 RF RE1 RL iE3 RE2 Feedback Network The forward amplifier in this circuit is NOT a two-port network because iE1 ≠ ii and iE3 ≠ io. However, However because iE1 and iE3 are closely related to ii and io, a two-port approach to feedback analysis can still be used. Begin g by y representing p g the feedback network as a two-port p network. B. Murmann EE214 Winter 2010-11 – Chapter 5 41 io Q3 RS + vS ~ – ii Q2 Q1 iE1 RC2 RC1 z11f + ~ z12f iE3 – RL iE3 z22f + z21f iE1 ~ – At the output, simply recognize that the feedback network “senses” iE3 rather than io, and that i iE3 = o α3 In effect, α3 is outside the feedback loop B. Murmann EE214 Winter 2010-11 – Chapter 5 42 At the input, input consider the following small-signal equivalent circuit RS + vS – ii + vπ1 – gm1vπ1 rπ1 iE1 z11f + z12f iE3 – v S = iiRS + v π1 + iE1z11f + z 12f iE3 ∴ v S − z 12f ⋅ io = i R + v π1 + iE1z11f α3 i S From this equation it is apparent that, although the z12f feedback voltage generator is in the emitter of Q1, it is still in series with, and subtracts from, the input voltage source, vS. Thus, the z12f generator can be moved f from the th emitter itt to t the th base b off Q1 B. Murmann EE214 Winter 2010-11 – Chapter 5 43 Next, N t to t determine d t i the th z parameters t for f the th feedback f db k network, t k consider id the th following circuit RF + v1 – i1 v z 12f = 1 i2 RE1 i1 =0 RE2 + v2 – ⎛ ⎛ RE1 ⎞ v 2 ⎜ =⎜ ⎟ ⎝ RE1 + RF ⎠ ⎜ i 2 ⎝ z 11f = i2 z 22f = v1 i1 = RE1 || (RF + RE2 ) i2 =0 v2 i2 = RE2 || (RF + RE1) i1 =0 ⎞ ⎛ RE1 ⎞ ⎟= z ⎟ ⎜⎝ RE1 + RF ⎟⎠ 22f i1 =0 ⎠ ⎛ RE1 ⎞ ⎡ RE2 (RE1 + RF ) ⎤ RE1 RE2 =⎜ ⎢ ⎥= ⎟ ⎝ RE1 + RF ⎠ ⎣ RE1 + RE2 + RF ⎦ RE1 + RE2 + RF z21f = z12f, but z21f is neglected in comparison with z21a B. Murmann EE214 Winter 2010-11 – Chapter 5 44 Based B d on th the preceding di results, lt th the series-series i i ttriple i l can b be modeled d l d with the following equivalent circuit io Q3 Q2 RS Q1 – ~ vfb=vf + + vS ~ – RE1 RC2 RC1 RF RF RE2 + vf – RL iE3 RE2 RE1 In this circuit we have managed to “break” the loop at the output of the feedback network. network Thus Thus, the circuit can be used to determine a a, f and T T. B. Murmann EE214 Winter 2010-11 – Chapter 5 45 Note that a and f for the series-series triple can be defined in terms of either io or iE3. If A= io i = α 3 E3 vS vS and a and f are defined with respect to io, then f= ⎞ z12f RE1RE2 1 ⎛ = ⎜ α 3 ⎝ RE1 + RE2 + RF ⎟⎠ α3 As T = af becomes large A≅ ⎛ R + R + RF ⎞ 1 = α3 ⎜ E1 E2 ⎟ f RE1RE2 ⎝ ⎠ Thus,, A is not desensitized to α3 by y the feedback loop p B. Murmann EE214 Winter 2010-11 – Chapter 5 46 Series-Shunt Feedback In a series series-shunt shunt feedback amplifier, amplifier the forward amplifier and feedback network share the same input current and output voltage. Therefore a hybrid h-parameter representation is used for the two-port networks. v1 = h11 i1 + h12 v 2 i2 = h21 i1 + h22 v 2 where h11 h21 B. Murmann v1 i1 i2 i1 h12 v 2 =0 h22 v 2 =0 v1 v2 i1 = 0 i2 v2 i1 = 0 EE214 Winter 2010-11 – Chapter 5 47 Series-Shunt Feedback, cont’d Thus, the two-port model for a series-shunt feedback circuit is Forward amplifier ii vS B. Murmann EE214 Winter 2010-11 – Chapter 5 48 Series-Shunt Feedback, cont’d Summing voltages at the input and currents at the output, v S = (zS + h11a + h11f )ii + (h12a + h12f ) v o 0 = (h21a + h21f )i i + (yL + h22a + h22f )io Define zi zS + h11a + h11f yo yL + h22a + h22f and neglect reverse transmission through the forward amplifier and feedforward through the feedback network; that is, assume h12a << h12f and h21a >> h21f B. Murmann EE214 Winter 2010-11 – Chapter 5 49 Series-Shunt Feedback Equations With the simplifying assumptions, the series-series feedback amplifier can be represented by a two-port model with ⎛ −h ⎞ 21a ⎜ ⎟ z y vo ⎝ i o ⎠ a A= = = vS ⎛ −h 1+ af h ⎞ 1+ ⎜ 21a ⎟ h12f ⎝ z i yo ⎠ where a=− h21a z i yo f = h12f B. Murmann EE214 Winter 2010-11 – Chapter 5 50 Series-Shunt Feedback Equivalent Two-Port Networks Equivalent E i l t ttwo-portt circuits i it can now b be id identified tifi d for f a and d f in i which hi h th the loading of the feedback network is included in the forward amplifier New forward amplifier ii yo zi + vo – vS B. Murmann EE214 Winter 2010-11 – Chapter 5 51 Shunt-Series Feedback In a shunt-series feedback circuit, circuit the forward amplifier and feedback network share the same input voltage and output current. Thus, a hybrid gparameter representation is used for the two-port networks. i1 = g11v1 + g12 i2 v 2 = g21v1 + g22 i2 where g11 i1 v1 i g12 v2 v1 g22 2 =0 g21 B. Murmann i2 =0 i1 i2 v1 = 0 v2 i2 EE214 Winter 2010-11 – Chapter 5 v1=0 52 Shunt-Series Feedback, cont’d The two-port model for a shunt-series feedback amplifier is thus Forward amplifier iS B. Murmann EE214 Winter 2010-11 – Chapter 5 53 Shunt-Series Feedback, cont’d Summing currents at the input and voltages at the output, iS= (yS + g11a + g11f )v i + (g12a + g12f )io 0 = (g21a + g21f )v i + (zL + g22a + g22f )io Define yi y S + g11a + g11f zo zL + g22a + g22f and neglect reverse transmission through the forward amplifier and feedforward through the feedback network; that is, assume g12a << g12f and g21a >> g21f B. Murmann EE214 Winter 2010-11 – Chapter 5 54 Shunt-Series Feedback Equations With the simplifying assumptions the shunt-series feedback circuit can described in the form of the ideal feedback equation ⎛ −g21a ⎞ ⎜ yz ⎟ i ⎝ i o ⎠ a A= o = = iS ⎛ −g 1+ af g ⎞ 1+ ⎜ 21a ⎟ g12f ⎝ yi zo ⎠ where a=− g21a y i zo f = g12f B. Murmann EE214 Winter 2010-11 – Chapter 5 55 Shunt-Series Feedback Equivalent Two-Port Networks Based on the above definitions and assumptions, equivalent two-port networks for a shunt-series amplifier can be defined as follows N New fforward d amplifier lifi zo + iS vi io yi – B. Murmann EE214 Winter 2010-11 – Chapter 5 56 Shunt-Series Feedback Pair io AC circuit: Q2 iS Q1 + vi – RS RC1 RF RL iE2 RE Feedback network The forward amplifier in this circuit is not a two-port network b because io ≠ iE2. However, H th the circuit i it can b be analyzed l d iin a ffashion hi similar to the series-series triple. The feedback network can be represented by a two-port characterized by g parameters. B. Murmann EE214 Winter 2010-11 – Chapter 5 57 io Q2 iS Q1 + vi – RS g11f RC1 RL iE2 g22f + ~ g21f vi – g12f iE2 The g parameters for the feedback network can be determined from the following circuit : i1 + v1 – B. Murmann ~ RF RE + v2 – EE214 Winter 2010-11 – Chapter 5 i2 58 g11f = i1 v1 g 22f = g12f = g21f = = i2 = 0 v2 i2 = RE || RF v1 = 0 i1 i2 1 RE + RF =− v1 = 0 v2 v1 RE RE + RF = − g12f i2 = 0 g21f is neglected in comparison with g21a The shunt-series p pair can then be redrawn as follows : B. Murmann EE214 Winter 2010-11 – Chapter 5 59 io Q2 iS + vi – Q1 RF+RE RC1 RS iF iE2 = –iiF RF RL iE2 RE A i we’ve Again, ’ managed d tto b break k th the lloop att th the output t t off th the ffeedback db k network. If a and f are defined in terms of io, then B. Murmann A= io i = α 2 E2 iS iS f= iF g ifb 1 ⎛ RE ⎞ = − = 12f = − io io α2 α 2 ⎜⎝ RE + RF ⎟⎠ EE214 Winter 2010-11 – Chapter 5 60 Chapter 6 Frequency Response of Feedback Amplifiers B. Murmann Stanford University y Reading Material: Section 9.1, 9.2, 9.3, 9.4, 9.5 Gain and Bandwidth Consider a feedback amplifier with a single pole in the response of the forward-path amplifier and feedback that is frequency independent. vi + – – a(s) vo vfb f a(s) = a0 1− s p1 A(s) v o (s) a(s) a(s) = = v i (s) 1+ a(s) ⋅ f 1+ T(s) where T(s) B. Murmann a(s) ⋅ f = "Loop Loop Gain Gain" EE214 Winter 2010-11 – Chapter 6 2 a0 a0 1− s p1 A(s) = = s a0 f 1− + a0 f 1+ p1 1− s p1 1 = a0 ⋅ 1+ a0 f = Ao ⋅ 1 1− s⎛ 1 ⎞ p1 ⎜⎝ 1+ a0 f ⎟⎠ 1 ⎡ ⎤ s 1− ⎢1 ⎥ ⎣ p1(1+ T0 ) ⎦ where A 0 = A(0) = a0 1+ a0 f T0 = T(0) = a(0) ⋅ f = a0 f = "Low Frequency Loop Gain" B. Murmann EE214 Winter 2010-11 – Chapter 6 3 Thus, feedback reduces the gain by (1+T0) and increases the –3dB bandwidth by (1+T0) for a “one-pole” forward-path amplifier. The Gain x Bandwidth (GBW) product remains constant. 20 log10 a0 20 log10 |a(jω)| 20 log10 (1+T0) 20 log10 A0 20 log10 |A(jω)| |p1| B. Murmann (1+T0)|p1| EE214 Winter 2010-11 – Chapter 6 4 Locus of the pole of A(s) in the s-plane: jω s plane s-plane x x (1+T0)p1 p1 σ Pole “moves” from p1 to (1+T0)p1 Note that ⎛a ⎞ 20 ⋅ log g10 a0 − 20 ⋅ log g10 A0 = 20 ⋅ log g10 ⎜ 0 ⎟ ⎝ A0 ⎠ = 20 ⋅ log10 (1+ T0 ) ≅ 20 ⋅ log g10 T0 B. Murmann when T0 >> 1 EE214 Winter 2010-11 – Chapter 6 5 At the frequency ω0 = (1+T0)|p1| a(jω 0 ) = A0 and therefore T(jω 0 ) = a(jω 0 ) ⋅ f = A 0 f = a0 f 1+ a0 f ≈1 Thus, ω0 is the unity-gain bandwidth of the loop gain, T(s). B. Murmann EE214 Winter 2010-11 – Chapter 6 6 Instability At a frequency where the phase shift around the loop of a feedback amplifier reaches ±180o the feedback becomes positive. In that case, pg gain is g greater than unity, y, the circuit is unstable. if the loop For a “single-pole” forward path amplifier stability is assured because the maximum phase shift is 90o. However, if a(s), or in general T(s), has multiple poles poles, the amount of loop gain that can be used is constrained. y of a feedback amplifier p can be assessed from: The stability – Nyquist diagram (polar plot of loop gain with frequency as a parameter) – Bode B d plot l t (plot ( l t off loop l gain i and d phase h as ffunctions ti off ffrequency)) – Locus of poles (root locus) of A(s) in the s-plane B. Murmann EE214 Winter 2010-11 – Chapter 6 7 Bode Stability Criterion A feedback system is unstable when |T(jω)| > 1 at the frequency where Phase[T(jω)] = -180° Phase margin – Defined at the frequency where |T(jω)| =1 1 Æ ω0 PM = 180° + Phase ⎡⎣T ( jω0 ) ⎤⎦ Gain margin – Defined at the frequency where Phase[T(jω)] = -180° Æ ω180 GM = B. Murmann 1 T ( jω180 ) EE214 Winter 2010-11 – Chapter 6 8 Example 20 log10 A0 Gain Margin ω0 | p2 | Phase Margin B. Murmann EE214 Winter 2010-11 – Chapter 6 9 Practical Phase Margins P Practical ti l circuits i it ttypically i ll use phase h margins i greater t 45° – For continuous time amplifiers, a common target is ~60° – For switched capacitor circuits, a phase margin of ~70° is desirable • See S EE315A In order to see the need for phase margin >45°, investigate the closedloop behavior of the circuit at ω = ω0 T( jω0 ) = a( jω0 ) ⋅ f = 1 A( jω0 ) = = B. Murmann ⇒ a( jω0 ) = 1 f (assuming f is real) a( jω0 ) 1 + a( jω0 ) a( jω0 ) a( jω0 ) 1+ e jφ[a( jω0 )] = a( jω0 ) 1 + e j(PM−180°) EE214 Winter 2010-11 – Chapter 6 10 PM = 45o a(jω 0 ) A(jω (jω 0 ) = = − j135° 1+ e a(jω 0 ) = 0.3 − 0.7j A(jω 0 ) = ∴ a(jω 0 ) = 0.76 a(jω 0 ) 1− 0.7 − 0.7j 1.3 ≅ 1.3A0 f Thus, |A(jω0)| “peaks” at ω = ω0, it’s nominal –3dB bandwidth. B. Murmann EE214 Winter 2010-11 – Chapter 6 11 PM = 60o A(jω 0 ) = a(jω 0 ) − j120° = 1+ e a(jω 0 ) = 0.5 − 0.87j ∴ A(jω 0 ) = a(jω 0 ) = a(jω 0 ) 1− 0.5 − 0.87j 1 ≅ A0 f PM = 90o A(jω 0 ) = ∴ B. Murmann A(jω 0 ) = a(jω 0 ) 1+ e − j90° a(jω 0 ) 1.4 = = a(jω 0 ) 1− j 0.7 ≅ 0.7A0 f EE214 Winter 2010-11 – Chapter 6 12 Closed-Loop Response for Various Phase Margins Text, p. 632 B. Murmann EE214 Winter 2010-11 – Chapter 6 13 Frequency Compensation (1) Frequency compensation refers f to the means by which the frequency f response of the loop gain in a feedback amplifier is altered so as to ensure adequate phase margin under the expected closed-loop conditions. Because operational amplifiers must often be compensated for use with a variety of feedback networks, compensation is usually accomplished by modifying only the forward path of the loop (i.e. the op-amp itself). In special i l purpose wideband id b d amplifier lifi d design, i th the response off th the feedback f db k network may also be altered. Several types yp of frequency q y compensation p are used in p practice,, e.g. g Narrowbanding (lag compensation) Feedforward (lead compensation) Pole splitting (Miller compensation, cascode compensation) Feedback (phantom) zero compensation B. Murmann EE214 Winter 2010-11 – Chapter 6 14 Narrowbanding (1) C Create t ad dominant i t pole l iin a(s) ( ) tto rollll off ff |T(j |T(jω)| )| att a frequency f low l enough to ensure adequate phase margin (f=1) Text, p. 634 B. Murmann EE214 Winter 2010-11 – Chapter 6 15 Narrowbanding (2) N Note t th thatt in i the th example l off the th previous i slide lid ((with ith ff=1, 1 and d PM PM=45°), 45°) the th closed-loop bandwidth is limited to approximately ωp1, the frequency of the closest non-dominant pole Thi This can b be acceptable t bl if ωp1 is i sufficiently ffi i tl llarge, as th the e.g. case ffor th the pole introduced by a cascode Consider e.g. the amplifier below – Cp introduces i t d a non-dominant d i t pole l att hi high h ffrequencies i – CL is adjusted until the circuit achieves the desired phase margin • Therefore, this type of narrowbanding is called “load compensation” B. Murmann EE214 Winter 2010-11 – Chapter 6 16 Two-Stage Amplifier But, what if we would like to stabilize an amplifier that has two poles at relatively low frequencies? Consider e.g. the two-stage amplifier shown below, and assume that ωp1=1/R1C1 and ωp2=1/R2C2 are comparable B. Murmann EE214 Winter 2010-11 – Chapter 6 17 Pole Splitting (1) R1 vi gm1 Cc C1 R2 vo gm2 C2 Purposely connect an additional capacitor between gate and drain of M2 (Cc = “compensation capacitor") Two interesting things happen – Low frequency input capacitance of second stage becomes large – exactly what we need for low ωp1 – At high frequencies, Cc turns M2 into a “diode connected device” – low impedance, p , i.e. large g ωp2 ! B. Murmann EE214 Winter 2010-11 – Chapter 6 18 Pole Splitting (2) F From the th generall CS/CE stage t analysis l i off Ch Chapter t 4 4, we can approximate resulting poles and zeros as follows – See also text, section 9.4.2 p1 ≅ − 1 gm2R2 R1CC p2 ≅ − gm2CC C1C2 + CC (C1 + C2 ) z=+ gm2 CC Increasing Cc reduces ωp1, and increases ωp2 − A very nice “knob” for adjusting the phase margin of the circuit The zero introduced due to Cc occurs at high frequencies and is not always troublesome In cases where the zero affects stability, y several options p exists to mitigate the problem − Nulling resistor, cascode compensation, etc. − See EE114, EE315A B. Murmann EE214 Winter 2010-11 – Chapter 6 19 Pole Splitting (3) Text, p. 642 c c c B. Murmann EE214 Winter 2010-11 – Chapter 6 20 Intuitive Derivation of Pole Split Using Shunt-Shunt Feedback a(s) = − gm2R2R1 (1 + sR1 [C1 + Cc ]) (1 + sR2 [C2 + Cc ]) Mag ( jω) f(s) = −sCf a(s) 1 f(s) ω B. Murmann EE214 Winter 2010-11 – Chapter 6 21 Nested Miller Compensation Text,, p. p 655 Mag ( jω) Cm2 Cm1 gm1, gm2 gm0, gm1, gm2 ω B. Murmann EE214 Winter 2010-11 – Chapter 6 22 Feedforward Compensation e.g. Thandri, JSSC 2/2003 |a(jω)| Parallel path through gm3 dominates transfer function at high frequencies and returns the circuit behavior back to first order |pd| ω |p1| |z1| φ(jω) ω – π/2 –π B. Murmann The doublet p1, z1 can make it difficult to achieve a fast transient response – See Kamath, JSSC 12/1974 EE214 Winter 2010-11 – Chapter 6 23 Feedback Zero Compensation Example: Mag ( jω) Due to the zero introduced in the feedback network network, T(j T(jω)) behaves like a first order system near unity crossover a(s) ( ) 1 f(s) ω B. Murmann Closed-loop bandwidth is approximately equal to the frequency of the zero in the feedback network – Can be far beyond second pole EE214 Winter 2010-11 – Chapter 6 24 Introduction to Root Locus Techniques: Three Pole Amplifier C Consider id a ffeedback db k network t k consisting i ti off a fforward d amplifier lifi with ith th three identical poles, and a feedback network with a constant transfer function f a0 3 a(s) = ⎛ s⎞ ⎜1− ⎟ p a0 a(s) 1⎠ ⎝ A(s) = = = 3 a0 1 + a(s)f ⎛ s⎞ 1+ a f 3 ⎜ 1 − ⎟ + T0 ⎛ s⎞ ⎝ p1 ⎠ − 1 ⎜ ⎟ p 1⎠ ⎝ a0 ⎛ s⎞ ⎜1− ⎟ ⎝ p1 ⎠ 3 T0 = a0 f The poles of A(s) are therefore the solution to 3 ⎛ s⎞ ⎜ 1 − ⎟ + T0 = 0 ⎝ p1 ⎠ B. Murmann EE214 Winter 2010-11 – Chapter 6 25 3 ⎛ s⎞ ⎜ 1 − ⎟ = −T0 ⎝ p1 ⎠ ⎛ s⎞ ⎜ 1 − ⎟ = 3 −T0 = − 3 T0 ⎝ p1 ⎠ ( = p (1 − = p (1 − 3 3 T0 e ( 3 T0 e j60° s1 = p1 1 + 3 T0 s2 s3 1 1 0 = 1 − Re ) T0 e j60° ) − j60° ) ⎛ ⎛ s⎞ s⎞ or ⎜ 1 − ⎟ = 3 T0 e j60° or ⎜ 1 − ⎟ = 3 T0 e− j60° ⎝ p1 ⎠ ⎝ p1 ⎠ “root locus” ) 0 = 1 − 3 T0 cos(60°) ⇒ T0 = 8 B. Murmann EE214 Winter 2010-11 – Chapter 6 26 Root-Locus Method We can generalize the above example to gain insight into the frequency response of any feedback amplifier by examining the “movement” of the closedloop poles in the s-plane as a function of the low-frequency loop gain, T0. Consider a generalized feedback amplifier with both a and f dependent on frequency. + vi – a(s) – vo vfb f(s) v o (s) a(s) a(s) = = v i (s) 1+ a(s)f(s) 1+ T(s) A(s) B. Murmann EE214 Winter 2010-11 – Chapter 6 27 The poles of A(s) are the roots of 1 + T(s) = 0 0. In general, a(s) = a0 1+ a1s + a 2s2 + ⋅ ⋅ ⋅ 2 1+ b1s + b 2s + ⋅ ⋅ ⋅ = a0 Na (s) Da (s) and f(s) = f0 1+ c1s + c 2s2 + ⋅ ⋅ ⋅ 2 1+ d1s + d2s + ⋅ ⋅ ⋅ = f0 Nf (s) D f (s) Thus, A(s) = a0 Na (s)D f (s) Da (s)D f (s) + T0 Na (s)Nf (s) where T0 = a0f0 . B. Murmann EE214 Winter 2010-11 – Chapter 6 28 The zeros of A(s) are the zeros of a(s) and the poles of f(s). Th poles The l off A(s) A( ) are the th roots t off : Da (s) D f (s) + T0 Na (s)Nf (s) = 0 As T0 increases from 0 to ∞, the poles of A(s) move in the s-plane from the poles of T(s) to the zeros of T(s). The Root Locus is the paths the roots of 1 + T(s) = 0 trace in the s-plane s plane as T0 varies from 0 to ∞. B. Murmann EE214 Winter 2010-11 – Chapter 6 29 The root-locus construction rules follow from the equation 1+ T(s) = 1+ T0 ⋅ or equivalently q y T0 ⋅ Na (s) ⋅Nf (s) =0 Da (s) ⋅D f (s) Na (s) ⋅Nf (s) = −1 Da (s) ⋅D f (s) Thus T0 ⋅ (1− s za1)(1− s za2 ) ⋅ ⋅ ⋅ (1− s z f1)(1− s z f 2 ) ⋅ ⋅ ⋅ = −1 ((1− s pa1)(1− )( s pa2 ) ⋅ ⋅ ⋅ ((1− s p f1)(1− )( s p f 2 ) ⋅ ⋅ ⋅ where za1,za2 ,... = zeros of a(s) z f1,zf2 ,... = zeros of f(s) pa1,pa2 ,... = poles of a(s) pf1,pf2 ,... = poles l off f(s) f( ) B. Murmann zeros of T(s) poles of T(s) EE214 Winter 2010-11 – Chapter 6 30 The above equation can be rewritten as ⎡ (−pa1)(−pa2 ) ⋅ ⋅ ⋅ (−p f1)(−p f 2 ) ⋅ ⋅ ⋅ ⎤ T0 ⋅ ⎢ ⎥ ⎣ (−za1)(−z a2 ) ⋅ ⋅ ⋅ (−z f1)(−z f 2 ) ⋅ ⋅ ⋅ ⎦ ⎡(s − za1)(s − za2 ) ⋅⋅⋅ (s − zf1)(s − zf 2 ) ⋅⋅⋅⎤ ×⎢ ⎥ = −1 ⎣(s − pa1)(s − pa2 ) ⋅⋅⋅ (s − pf1)(s − pf 2 ) ⋅⋅⋅⎦ Expect all poles of T(s) to be in the left half plane (LHP). If all zeros of T(s) are in the LHP, or if there are an even number of zeros in the RHP RHP, then the first bracketed term in the above equation is positive, in which case ⎡ p p ⋅ ⋅ ⋅ p f1 p f 2 ⋅ ⋅ ⋅ ⎤ T0 ⋅ ⎢ a1 a2 ⎥ ⎢⎣ z a1 z a2 ⋅ ⋅ ⋅ z f1 z f 2 ⋅ ⋅ ⋅ ⎥⎦ ⎡(s − za1)(s − za2 ) ⋅⋅⋅ (s − zf1)(s − zf 2 ) ⋅⋅⋅⎤ ×⎢ ⎥ = −1 ⎣(s − pa1)(s − pa2 ) ⋅⋅⋅ (s − pf1)(s − pf 2 ) ⋅⋅⋅⎦ B. Murmann EE214 Winter 2010-11 – Chapter 6 31 Values of s satisfying the above equation are the poles of A(s) A(s). These values simultaneously fulfill both a phase condition and a magnitude condition, and these conditions define the points of the root locus. Phase Condition ⎡⎣ ∠(s − za1) + ∠(s − za2 ) + ⋅ ⋅ ⋅ + ∠(s − z f1) + ∠(s − z f 2 ) + ⋅ ⋅ ⋅⎤⎦ − ⎡⎣ ∠(s − pa1) + ∠(s − pa2 ) + ⋅ ⋅ ⋅ + ∠(s − p f1) + ∠(s − p f 2 ) + ⋅ ⋅ ⋅⎤⎦ = (2n − 1)π Magnitude Condition ⎡ p p ⋅ ⋅ ⋅ p f1 p f 2 ⋅ ⋅ ⋅ ⎤ ⎡ s − z a1 s − z a2 ⋅ ⋅ ⋅ s − z f1 s − z f 2 ⋅ ⋅ ⋅ ⎤ T0 ⋅ ⎢ a1 a2 ⎥ ⋅⎢ ⎥ = +1 ⎢⎣ z a1 z a2 ⋅ ⋅ ⋅ z f1 z f 2 ⋅ ⋅ ⋅ ⎥⎦ ⎢⎣ s − pa1 s − pa2 ⋅ ⋅ ⋅ s − p f1 s − p f 2 ⋅ ⋅ ⋅ ⎥⎦ B. Murmann EE214 Winter 2010-11 – Chapter 6 32 For the case where there are an odd number of zeros in the RHP, the magnitude condition remains the same as above, but the phase condition is changed. Specifically, ⎡ p p ⋅ ⋅ ⋅ p f1 p f 2 ⋅ ⋅ ⋅ ⎤ ⎡ s − z a1 s − z a2 ⋅ ⋅ ⋅ s − z f1 s − z f 2 ⋅ ⋅ ⋅ ⎤ T0 ⋅ ⎢ a1 a2 ⎥ ⋅⎢ ⎥ = +1 ⎢⎣ z a1 z a2 ⋅ ⋅ ⋅ z f1 z f 2 ⋅ ⋅ ⋅ ⎥⎦ ⎣⎢ s − pa1 s − pa2 ⋅ ⋅ ⋅ s − p f1 s − p f 2 ⋅ ⋅ ⋅ ⎥⎦ and ⎡⎣ ∠(s − za1) + ∠(s − za2 ) + ⋅ ⋅ ⋅ + ∠(s − z f1) + ∠(s − z f 2 ) + ⋅ ⋅ ⋅⎤⎦ − ⎡⎣ ∠(s − pa1) + ∠(s − pa2 ) + ⋅ ⋅ ⋅ + ∠(s − p f1) + ∠(s − p f 2 ) + ⋅ ⋅ ⋅⎤⎦ = 2nπ B. Murmann EE214 Winter 2010-11 – Chapter 6 33 The rules for constructing the root locus are based on the phase condition. The magnitude condition determines where, for a given T0, the poles of A(s) actually lie on along the locus. To determine if a point X in the s-plane lies on the root locus, draw vectors from the poles and zeros of T(s) to the point X. The angles of these vectors are then used to check the phase condition. B. Murmann EE214 Winter 2010-11 – Chapter 6 34 Root-Locus Construction Rules Rule 1: Branches of the root locus start at the poles of T(s), where T0 = 0, 0 and terminate on the zeros of T(s) T(s), where T0 = ∞. If T(s) has more poles than zeros, some branches terminate at infinity. Rule 2: The root locus is located along the real axis whenever there is an odd number of poles and zeros of T(s) between that portion of the real axis and the origin of the s-plane. Rule 3: All segments of the locus on the real axis between pairs of poles, or pairs of zeros, must branch out from the real axis axis. B. Murmann EE214 Winter 2010-11 – Chapter 6 35 Rule 4: Locus is symmetric with respect to the real axis because complex roots occur only in conjugate pairs. Rule 5: Branches leaving the real axis do so at right angles to it. Rule 6: Branches break away from the real axis at points where the vector sum of reciprocals of distances to the poles of ( ) equals q the vector sum of reciprocals p of distances to T(s) the zeros of T(s). Rule 7: Branches B h th thatt terminate t i t att infinity i fi it do d so asymptotically t ti ll tto straight lines with angles to the real axis of (2n –1)π/(Np – Nz), where Np = # of poles of T(s) and Nz = # of zeros of T(s) T(s). B. Murmann EE214 Winter 2010-11 – Chapter 6 36 R l 8: Rule 8 The asymptotes of the branches terminating at infinity all intersect the real axis at a single point given by σa = B. Murmann [ ] [ ] Σ poles of T(s) − Σ zeros of T(s) Np − Nz EE214 Winter 2010-11 – Chapter 6 37 2-Pole Example jω p2 B. Murmann p1 EE214 Winter 2010-11 – Chapter 6 σ 38 3-Pole Example Assume T(s) = T0 (1− s / p1)(1− s / p 2 )(1− s / p3 ) where p1 = −1× 106 sec−1 p2 = −2 × 106 sec−1 p3 = −4 × 106 sec−1 B. Murmann EE214 Winter 2010-11 – Chapter 6 39 3-Pole Example, cont’d σa σi B. Murmann EE214 Winter 2010-11 – Chapter 6 40 Breakaway Point, σi From Rule 6 1 1 1 + + =0 σi + 1 σi + 2 σi + 4 2 ∴ 3σ 3 i + 14σ 14 i + 14 = 0 2 ⎛ 7⎞ 7 14 1 σi = − ± ⎜ ⎟ − = − 7± 7 3 3 3 ⎝ 3⎠ ( ) = −3.22 or − 1.45 Since σi lies bet between een p1 and p2, σi = −1.45 × 106 sec−1 B. Murmann EE214 Winter 2010-11 – Chapter 6 41 Asymptote Intercept, σa From Rule 8 σa = (−1− 2 − 4) − 0 = −2.33 2 33 × 106 sec−1 3 To find the loop gain, T0, at which the poles moving from p1 and p2 b become complex, l substitute b tit t th the b breakaway k point, i t s = σi = –1.45, 1 45 into the magnitude condition and solve for T0. T0 p1 p 2 p3 s − p1 s − p 2 s − p3 ∴ T0 = =1 −1 1.45 45 + 1 −1 1.45 45 + 2 −1 1.45 45 + 3 (1)(2)(4) (0.45)(0.55)(2.55) = = 0.08 8 B. Murmann EE214 Winter 2010-11 – Chapter 6 42 The complex poles enter the RHP at approximately the points where their asymptotes cross the jω axis. At these points s = ± j(2.33) j(2 33) tan(60 ) = ±4j Substituting into the magnitude condition and solving for T0 T0 = = s − p1 s − p2 s − p3 p1 p2 p3 4j + 1 4j + 2 4j + 4 ( = = (1)(2)(4) 42 + 1 )( 4 2 + 22 8 )( 42 + 42 )= 17 ⋅ 20 ⋅ 32 8 (4.1)(4.5)(5.7) = 13.2 8 B. Murmann EE214 Winter 2010-11 – Chapter 6 43 Plotting the Root Locus in Matlab % root locus example s = tf('s'); p1=-1; p2=-2; p3=-4; a = 1 / [(1-s/p1)*(1-s/p2)*(1-/p3)] rlocus(a) Root Locus 10 8 6 Imaginary A Axis 4 2 0 -2 -4 -6 -8 -10 -14 14 -12 12 -10 10 -8 8 -6 6 -4 4 -2 2 0 2 4 Real Axis B. Murmann EE214 Winter 2010-11 – Chapter 6 44 3 Poles and 1 Zero jω p2 σ p1 Introducing a zero in T(s) can be used to stabilize a feedback amplifier. The use of zeros in the feedback path is an important aspect of wideband amplifier design. B. Murmann EE214 Winter 2010-11 – Chapter 6 45 Matlab Plot % root locus example s = tf('s'); z=-5; p1=-1; p2=-2; p3=-4; a = (1-s/z) / [(1-s/p1)*(1-s/p2)*(1-s/p3)] rlocus(a) ocus(a) Root Locus 15 10 Imaginary A Axis 5 0 -5 -10 -15 -6 6 -5 5 -4 4 -3 3 -2 2 -1 1 0 1 Real Axis B. Murmann EE214 Winter 2010-11 – Chapter 6 46 Chapter 7 Wideband Amplifiers B. Murmann Stanford University Reading Material: Section 9.5.3, 9.5.4 Overview How to build amplifiers with large gain and bandwidth? Options we’ll look at – Cascade of first order amplifier stages – Multi-stage Multi stage amplifiers with global feedback – Multi-stage amplifiers with local feedback B. Murmann EE214 Winter 2010-11 – Chapter 7 2 Cascade of N First-Order Stages Consider a cascade of N identical stages, each with the singlepole response A(jω) = A0 1+ jω ωB GBW = A 0ωB If there is no interaction among the stages, then the overall transfer function of the cascade is ⎛ ⎞ A0 AN (jω) = ⎜ ⎟ ⎝ 1+ jω ωB ⎠ N In this case A 0N = AN0 B. Murmann EE214 Winter 2010-11 – Chapter 7 3 Th –3dB The 3dB b bandwidth d idth off th the cascade d iis the th frequency, f ωBN, att which hi h AN (jωBN ) = Thus, A 0N 2 ⎡ A0 ⎢ ⎢ 1+ (ω ω B )2 BN ⎣ N ⎤ N ⎥ = A0 ⎥ 2 ⎦ 2 1+ (ωBN ωB ) = 21 N ωBN = ωB ⋅ 21 N − 1 B. Murmann GBWN = A N0 ωB ⋅ 21 N − 1 < A N0 ωB EE214 Winter 2010-11 – Chapter 7 4 Bandwidth Shrinkage 1 0.8 1 N 0.6 2 −1 0.4 0.2 0 2 4 6 8 10 N B. Murmann EE214 Winter 2010-11 – Chapter 7 5 Ideal Lowpass Response Suppose each stage in the cascade of N identical stages had an ideal lowpass response; that is, a magnitude response with an abrupt band edge transition between the passband and the stopband: |A(jω)| A0 ⇒ GBWN = A N0 ωB ωB ω In the design of wideband amplifiers it is therefore often desirable to h have a much h sharper h b band-edge d d ttransition iti than th iis obtained bt i d with ith a cascade of identical single-pole stages. B. Murmann EE214 Winter 2010-11 – Chapter 7 6 Maximally Flat Magnitude (MFM) Response An approximation to the ideal lowpass response that is commonly used in broadband amplifier design is the Maximally Flat Magnitude (MFM), or Butterworth, response. response The MFM response provides a magnitude (gain) that is flat over as much of the passband as possible and decreases monotonically with frequency (i no peaking). (i.e. ki ) It iis obtained bt i d b by setting tti as many d derivatives i ti off th the magnitude with respect to frequency as possible to zero at ω = 0. For an nth-order MFM response, dk H(jω) dω k B. Murmann =0 for 1 ≤ k ≤ 2n − 1 ω =0 EE214 Winter 2010-11 – Chapter 7 7 Th resulting The lti magnitude it d response iis H(jω) = 1 1+ (ω ( B)2n where n is the order of the response, B is the bandwidth of the ideal p response, p , and the magnitude g has been normalized to unity y lowpass at ω = 0. |H(jω)| 1 n=2 n=1 n=3 B B. Murmann EE214 Winter 2010-11 – Chapter 7 ω 8 There is no peaking in the MFM response, and as n increases the band edge transition becomes sharper. In the limit as n t ∞, the MFM response approaches the ideal lowpass response. To determine the location of the poles of H(s), note that 2 H(jω) = H(jω) ⋅H(− jω) = H(s) ⋅H(−s) s= jω Thus,, for an MFM response p H(s) ⋅H(−s) = = B. Murmann 1 1+ (s jB)2n = j2n j2n + (s B)2n (−1)n (−1)n + (s B)2n EE214 Winter 2010-11 – Chapter 7 9 The poles of H(s)×H(–s) are the 2n roots satisfying ⎛ s⎞ ⎜⎝ B ⎟⎠ 2n = −(−1)n = (−1)n+1 Th These roots t lie li equally ll spaced d on a circle i l in i th the s-plane l centered t d att the origin with radius B. The LHP roots are taken to be the poles of H(s), while those in the RHP are regarded d d as th the poles l off H( H(–s). ) B. Murmann EE214 Winter 2010-11 – Chapter 7 10 MFM Pole Locations n=1 jω p1 = − B σ –B n=2 pi = Be j3π 4 , Be j5π 4 jω B 45º σ B. Murmann EE214 Winter 2010-11 – Chapter 7 11 n=3 pi = Be j2π 3 , Be jπ , Be j4π 3 jω 60º σ B. Murmann EE214 Winter 2010-11 – Chapter 7 12 MFM Transient Response A potential disadvantage of the MFM response is that there is an overshoot in the step response when n ≥ 2. This may be an important consideration in applications such as pulse amplifiers. For a step input at t=0 n=3 3 Vo(t) n=1 n=2 t n = 2 Æ 4% overshoot n = 3 Æ 8% overshoot B. Murmann EE214 Winter 2010-11 – Chapter 7 13 Comparison of 2-Pole Responses Angle of Poles –3dB BW Overshoot 10-90% Rise Time Coincident poles 0° 0.64B 0 3.4/B MFM (Butterworth) 45° B 4% 2.2/B Chebyshev (1-dB ripple) 60° 1.3B 16% 1.6/B Bessel 30° 30 0 8B 0.8B 0 4% 0.4% 2 7/B 2.7/B B. Murmann EE214 Winter 2010-11 – Chapter 7 More in EE315A 14 Wideband Amplification Using Multi-Stage Feedback At most three gain stages can be effectively used within a single feedback loop. Therefore, there are four basic configurations. + + vi – + vo – ~ RF RE A V0 = Series-Shunt Pair B. Murmann v o RF ≈ v i RE EE214 Winter 2010-11 – Chapter 7 15 + io ii AI0 = RF i o RF ≈ ii RE RE Shunt-Series Pair B. Murmann EE214 Winter 2010-11 – Chapter 7 16 + io + vi – ~ RF RE1 A0 = RE2 i o RF + RE1 + RE2 ≈ vi RE1 ⋅RE2 Series-Series Triple B. Murmann EE214 Winter 2010-11 – Chapter 7 17 + + vo – ii RF A0 = vo ≈ RF ii Shunt-Shunt Triple B. Murmann EE214 Winter 2010-11 – Chapter 7 18 Analysis of Shunt-Series Pair + RC2 RC1 io Q2 Q1 ZL ii RF RE Assume ZL ≈ 0 and RS ≈ ∞ B. Murmann EE214 Winter 2010-11 – Chapter 7 19 Application Example: Optical Fiber Links (1) K. Ohhata et al., “A Wide-Dynamic-Range, HighTransimpedance Si Bipolar Preamplifier IC for 10-Gb/s Optical Fiber Links,” JSSC 1/1999. B. Murmann EE214 Winter 2010-11 – Chapter 7 20 Circuit Implementation Ohhata, JSSC 1/1999 B. Murmann EE214 Winter 2010-11 – Chapter 7 21 Small-signal equivalent circuit for evaluating the open-loop response of the shunt-series pair: io RC1 Q2 Q1 ii ifb iE2 RF RE B. Murmann ifb RF EE214 Winter 2010-11 – Chapter 7 RE 22 Note that in BJT amplifiers with a series output stage, the feedback network does not sample the output, io, directly, but rather the emitter current of Q2, iE2. Thus, a(s) and f(s) are defined as follows: a(s) iE2 ii f(s) ifb iE2 For these definitions A(s) io ⎛ a(s) ⎞ = −α 2 ⎜ ii ⎝ 1+ a(s)f(s) ⎟⎠ where α2 is the common-base current gain of Q2 B. Murmann EE214 Winter 2010-11 – Chapter 7 23 The feedback response response, f(s) f(s), for the shunt-series pair (w/o compensation) is simply f(s) = ifb RE =− iE2 RE + RF To estimate the forward p path response, p a(s), ( ) begin g with the following small-signal equivalent circuit: Cμ1 ii + R1 v1 – Cπ1 Cμ2 + RC1 v rπ2 2 gm1v1 – + vb – R1 = (RF + RE ) || rπ1 Cπ2 RE* io gm2vb iE2 RE* = RE || RF B. Murmann EE214 Winter 2010-11 – Chapter 7 24 First consider the transmission from the base to the emitter of the second stage iE2 = (v 2 − iE2RE* )(gm22 + Yπ 2 ) where Yπ22 gπ22 + sC π22 Thus, iE2 ⎡1+ 1 RE* (g ( m2 + Yπ 2 ) ⎤ = v 2 (g ( m2 + Yπ 2 ) ⎣ ⎦ ∴ B. Murmann gm2 + Yπ 2 iE2 = v 2 1+ RE* (gm2 + Yπ 2 ) EE214 Winter 2010-11 – Chapter 7 gm2 + Yπ 2 = gm2 + 25 gm2 + sC π 2 ≅ gm2 + sC π 2 β0 ⎛ C ⎞ = gm2 ⎜ 1+ s π 2 ⎟ = gm2 1+ sτ T2 gm2 ⎠ ⎝ ( ) Therefore, Therefore gm2 (1+ sτ T2 ) iE2 = v 2 1+ gm2RE* + sRE* C π2 ⎡ ⎤ ⎡ ⎤ ⎢ ⎥ ⎢ ⎥ ⎛ g ⎞⎢ ⎛ g ⎞⎢ ⎥ 1+ sτ 1+ sτ ⎥ m2 T2 m2 T2 =⎜ ⎟⎢ ⎥= ⎜ ⎥ * ⎟ ⎢ * * * ⎛ R C ⎞ ⎥ ⎝ 1+ gm2RE ⎠ ⎢ ⎛ g R ⎞⎥ ⎝ 1+ gm2RE ⎠ ⎢ E π2 m2 E 1+ sτ T2 ⎜ * ⎟ ⎢ 1+ s ⎜ 1+ g R* ⎟ ⎥ ⎢ ⎝ 1+ gm2RE ⎠ ⎦⎥ ⎝ m2 E ⎠ ⎦ ⎣ ⎣ B. Murmann EE214 Winter 2010-11 – Chapter 7 26 jω –ωT2 σ ⎛ 1+ g R* ⎞ m2 E −ω T2 ⎜ * ⎟ ⎝ gm2RE ⎠ Thus, the second stage is very broadband, and the pole-zero pair associated with it can usually be neglected. In that case, gm2 iE2 ≅ v 2 1+ gm2RE* gm2eq Also,, io = − α 2iE2 ≅ − iE2 B. Murmann EE214 Winter 2010-11 – Chapter 7 27 The main influence of the series feedback stage on a(s) is the loading on Q1. We can model the loading using the following approximation for the input impedance of the series feedback stage: + v2 – rπ2eq Cμ2 Cπ2eq where rπ2eq = rπ2 (1+ gm2RE* ) ≅ rπ2gm2RE* = β02RE* C π2eq = B. Murmann C π2 1+ gm2RE* ≅ C π2 ⎛ 1 ⎞ τ T2 ⎜ ⎟= gm2 ⎝ RE* ⎠ RE* EE214 Winter 2010-11 – Chapter 7 28 Thus, the equivalent circuit for determining the open-loop response v2(s)/ii(s) is simply: Cμ1 ii + R1 v1 – Cπ1 gm1v1 RL1 CL1 + v2 – where R1 = rπ1 || (RF + RE ) RL1 = RC1 || rπ2eq ≅ RC1 || β02RE* CL1 = Ccs1 + Cμ 2 + C π 2eq ≅ Ccs1 + Cμ 2 + B. Murmann a(s) we find RE* EE214 Winter 2010-11 – Chapter 7 Since and τ T2 29 iE2 ⎛ iE2 ⎞ ⎛ v 2 ⎞ =⎜ ⎟⎜ ⎟ ii ⎝ v 2 ⎠ ⎝ ii ⎠ gm2 iE2 1 ≅ gm2eq = ≅ * 2 * v2 1+ gm2RE RE ( ) ⎡ ⎛ R ⎞ ⎤ ⎡ 1− s z1 ⎤ ⎥⎢ a(s) = − ⎢gm1R1 ⎜ L1 ⎥ * ⎟ 2 ⎢⎣ ⎝ RE ⎠ ⎥⎦ ⎢⎣ 1+ b1s + b 2s ⎥⎦ Where (see Chapter 4) b1 = R1(Cπ1 + Cμ1) + RL1(CL1 + Cμ1) + gm1R1RL1Cμ1 b 2 = R1RL1(C π1CL1 + C π1Cμ1 + CL1Cμ1) z1 = + B. Murmann gm1 C μ1 EE214 Winter 2010-11 – Chapter 7 30 If a dominant pole condition exists, with |p1| << |p2|, then p1 ≅ − =− 1 b1 1 R1C π1 + RL1CL1 + gm1R1RL1Cμ1 and p2 = R1C π1 + RL1CL1 + gm1R1RL1Cμ1 1 ≅− b 2p1 R1C π1RL1CL1 ⎛ 1 gm1Cμ1 ⎞ 1 = −⎜ + + ⎟ ⎝ R1C π1 RL1CL1 C π1CL1 ⎠ B. Murmann EE214 Winter 2010-11 – Chapter 7 In the s-plane 31 jω p2 p1 σ Unfortunately, non-dominant poles (and RHP zeros) of T(s) tend to push the poles of A(s) toward the RHP B. Murmann EE214 Winter 2010-11 – Chapter 7 32 Feedback (Phantom) Zeros The flat bandwidth and loop gain achievable in a shunt-series pair can be increased significantly by introducing a feedback (phantom) zero. This Thi is i accomplished li h d by b iincluding l di a capacitor, it CF, in i shunt h t with ith RF; the feedback network then becomes: CF ifb RF B. Murmann RE iE2 EE214 Winter 2010-11 – Chapter 7 33 Neglecting the loading of CF at the input and the emitter of Q2 in the forward path (since it appears in shunt with RE), a(s) remains the same as w/o CF. However, ⎛ RE ⎞ ⎛ 1 − s zF ⎞ f(s) = − ⎜ ⎟⎜ ⎟ ⎝ RE + RF ⎠ ⎝ 1 − s pF ⎠ where zF = − 1 RFCF pF = − R + RE 1 =− F (RF || RE )CF RFRECF Since A 0 ≅ (RF + RE ) RE pF = A0 zF B. Murmann EE214 Winter 2010-11 – Chapter 7 34 Since CF is chosen so that |zF| is on the order of the bandwidth of A(s), pF can be ignored unless the low-frequency closed loop gain, A0, is small. Thus, ⎛ RE ⎞ ( ) ≅ −⎜ f(s) ⎟ ((1− sRFCF ) ⎝ RE + RF ⎠ CF is chosen so as to obtain the desired closed-loop pole positions. For example, p p , for an MFM response: p jω σi zF B. Murmann σ p2 45º p1 EE214 Winter 2010-11 – Chapter 7 35 If zF is reasonably remote from p1 and p2, then σi = p1 + p 2 2 The root locus is a circle of radius |zF – σi| centered at zF. For each branch of the locus there are two intersections with the two lines at 45° to the negative real axis that correspond to an MFM response 45 response. If |zF| >> |σi|, then for an MFM response B = 2 zF and the closed-loop pp pole p positions,, s1 and s2, are s1,s2 ≅ zF (−1± j) = B. Murmann B 2 EE214 Winter 2010-11 – Chapter 7 (−1± j) 36 The loop gain, T0, needed to place the closed-loop poles at these positions is T0 = = B. Murmann s1 − p1 s2 − p 2 p1 p 2 2 zF 2 p1 p 2 = = B2 p1 p 2 2R1C π1RL1CL1 (RFCF )2 EE214 Winter 2010-11 – Chapter 7 37 Broadband Amplifier Design with Local Feedback Rein & Moller, JSSC 8/1996 In a cascade of amplifier stages, the interaction (loading) between adjacent stages can be reduced by alternating local series and shunt feedback circuits. B. Murmann EE214 Winter 2010-11 – Chapter 7 38 Advantages & Disadvantages of Local Feedback Cascades Advantages • No N iinstability bili • Lower gain sensitivity to component and device parameter variations than amplifiers without local feedback Disadvantages • Hi Higher h gain i sensitivity iti it (less (l lloop gain) i ) th than multi-stage lti t feedback amplifiers (i.e. multi-stage amplifiers with global feedback) • Bandwidth obtainable is typically slightly less than that possible in multi-stage feedback amplifiers B. Murmann EE214 Winter 2010-11 – Chapter 7 39 Application Example (1) Poulton, ISSCC 2003 B. Murmann EE214 Winter 2010-11 – Chapter 7 40 Application Example (2) Poulton, ISSCC 2003 B. Murmann EE214 Winter 2010-11 – Chapter 7 41 Application Example (3) Poulton, ISSCC 2003 BW ~ 6 GHz B. Murmann EE214 Winter 2010-11 – Chapter 7 42 Cherry-Hooper Amplifier + RF RE Shunt F/B Series F/B Stage Stage E. Cherry and D. Hooper, Proc. IEE, Feb. 1963 B. Murmann EE214 Winter 2010-11 – Chapter 7 43 Analysis A(0) = vo ⎛ 1 ⎞ RF ≅ ⎜− ⎟ ( −RF ) = v in ⎝ RE ⎠ RE Thus, the low-frequency gain is approximately equal to the ratio of the p and two feedback resistors,, and is therefore insensitive to component device parameter values While the series feedback stage usually has a large bandwidth, there is p and output p nodes of the considerable interaction between the input shunt feedback stage. However this interaction can be controlled and can be used to position the poles to achieve a particular response, such as MFM. To see how the pole positions in a local feedback cascade can be controlled, we first consider the local feedback stages individually B. Murmann EE214 Winter 2010-11 – Chapter 7 44 Series Feedback Stage Small-signal equivalent circuit: RS vi + v1 – ~ io Cμ rπ gmv1 Cπ + vE – ZL RE Include transistor rb in RS Neglect rμ, ro, re, and rc Assume that ZL ≈ 0; then there is no Milller effect and Cμ is just a (small) capacitance to ground which often has negligible impact. B. Murmann EE214 Winter 2010-11 – Chapter 7 45 Define Yπ 1 + sC π rπ Then v i − (v1 + vE ) = v1Yπ RS v1Yπ + gmv1 = vE RE Substituting for vE in the first of these two equations v i − v1 ⎡⎣1+ (gm + Yπ ) RE ⎤⎦ = v1YπRS ∴ B. Murmann v1 1 = v i 1+ (gm + Yπ ) RE + YπRS EE214 Winter 2010-11 – Chapter 7 46 Since io = –gmv1 io gm =− vi 1+ gmRE + (RS + RE )Yπ ⎡ ⎤ ⎢ ⎥ ⎛ gm ⎞ ⎢ 1 ⎥ = −⎜ ⎟ ⎢ ⎛ R +R ⎞ ⎛ ⎥ 1+ g R ⎞ ⎝ 1 m E⎠ E ⎢ 1+ ⎜ S ⎥ + sC π⎟ ⎟⎜ ⎢⎣ ⎝ 1+ gmRE ⎠ ⎝ rπ ⎠ ⎥⎦ ⎡ ⎤ ⎥ ⎢ ⎛ gm ⎞ ⎢ 1 ⎥ = −⎜ ⎟ ⎢ ⎛ R + RE ⎞ ⎛ ⎞ ⎛ RS + RE ⎞ ⎥ ⎝ 1+ gmRE ⎠ 1 ⎢ 1+ + sC 1 ⎜ S C ⎟⎜ π⎜ ⎟ ⎟⎥ ⎢⎣ ⎝ rπ ⎠ ⎝ 1+ gmRE ⎠ ⎝ 1+ gmRE ⎠ ⎦⎥ B. Murmann EE214 Winter 2010-11 – Chapter 7 47 Usually, Usually ⎛ RS + RE ⎞ ⎛ ⎞ gm (RS + RE ) ⎛ ⎞ 1 1 ⎜ r ⎟ ⎜ 1+ g R ⎟ = ⎜ ⎟ β0 ⎝ 1+ gmRE ⎠ ⎝ ⎠⎝ π m E⎠ << 1 Then ⎛ io 1 ⎞ ≅ −gmeq ⎜ ⎟ vi ⎝ 1− s p1 ⎠ where gmeq = gm 1+ gmRE ⎛ 1+ gmRE ⎞ ⎛ 1 ⎞ p1 = − ⎜ ⎟⎜ ⎟ ⎝ C π ⎠ ⎝ RE + RS ⎠ B. Murmann EE214 Winter 2010-11 – Chapter 7 48 If gmRE >> 1 gmeq ≅ 1 RE and p1 ≅ − ⎛ Cπ + Cμ ⎞⎛ RE ⎞ ⎛ RE ⎞ gm ⎛ RE ⎞ ⎜ ⎟ = − ωT ⎜ ⎟⎜ ⎟ ≅ − ωT ⎜ ⎟ Cπ ⎝ RE + RS ⎠ ⎝ Cπ ⎠⎝ RE + RS ⎠ ⎝ RE + RS ⎠ Note that this result corresponds to the dominant time constant found via ZVTC analysis in handout 7. Thus, for RE >> RS, p1 → – ωT. In some cases, the bandwidth may then be actually limited by Cμ (which was neglected in this analysis). B. Murmann EE214 Winter 2010-11 – Chapter 7 49 Emitter Peaking The bandwidth of the series feedback stage can be increased by introducing an “emitter peaking” capacitor in shunt with RE. io RS + vi – Q1 ~ RE CE N a large Not l b bypass capacitor To analyze this circuit, substitute ZE for RE in the preceding analysis where analysis, ZE = B. Murmann RE 1 = YE 1+ sRECE EE214 Winter 2010-11 – Chapter 7 50 Then, defining τE RECE and assuming τ T ≅ io =− vi gm 1+ gmZE + (RS + ZE )( 1 + sC π ) rπ gm =− 1+ − Cπ gm ⎞ ⎛ RE ⎞ ⎛ RS 1 gm + + sC π ⎟ + sC πRS + ⎜ ⎜ ⎟ rπ rπ ⎝ 1+ sτE ⎠ ⎝ ⎠ gm ⎛ g R ⎞⎛ R C ⎞ 1+ S + sC πRS + ⎜ m E ⎟ ⎜ 1+ s π ⎟ rπ gm ⎠ ⎝ 1+ sτE ⎠ ⎝ since gm >> B. Murmann 1 rπ EE214 Winter 2010-11 – Chapter 7 ∴ io ≅− vi 51 gm 1+ ⎛ 1+ sτ T ⎞ RS + sC πRS + gmRE ⎜ ⎟ rπ ⎝ 1+ sτE ⎠ If CE is chosen so that τE = τ T and it is true that gmRE >> Then B. Murmann RS gmRS = rπ β0 ⎡ ⎤ ⎢ ⎥ ⎛ gm ⎞ ⎢ io 1 ⎥ ≅ −⎜ ⎟⎢ ⎥ vi 1+ g R ⎛ ⎞ RS ⎝ m E⎠ ⎢ 1+ sC π ⎜ ⎟⎥ ⎝ 1+ gmRE ⎠ ⎥⎦ ⎢⎣ EE214 Winter 2010-11 – Chapter 7 52 Th result The lt is i a single-pole i l l response with ith p1 = − ⎛R ⎞ 1+ gmRE ≅ −ω T ⎜ E ⎟ RSC π ⎝ RS ⎠ In this case, if RE > RS, then |p1| > ωT (!) B. Murmann EE214 Winter 2010-11 – Chapter 7 53 Note that τE > τT results in a pole-zero separation that can cause peaking in the frequency response jjω p2 p1 σ z1 |io/vi| |z1| |p1| B. Murmann |p2| EE214 Winter 2010-11 – Chapter 7 ω 54 Shunt Feedback Stage Small-signal equivalent circuit CF RF + v1 – ii rπ Cπ gmv1 RL CL + vo – Include transistor Cμ in CF Neglect RS or include it in rπ Neglect rb, rc, re, and rμ Include ro and Ccs in RL and CL B. Murmann EE214 Winter 2010-11 – Chapter 7 55 D fi Define τF RFCF τL RLCL and B. Murmann YF = 1 1 1 + sCF = (1+ sCFRF ) = (1+ sτF ) RF RF RF YL = 1 1 + sCL = ((1+ sτL ) RL RL Yπ = 1 + sC π rπ EE214 Winter 2010-11 – Chapter 7 56 From the equivalent circuit: ii + (v o − v1)YF = v1Yπ (1) v o YL + gmv1 + (v o − v1)YF = 0 (2) Then from (2) ⎛ Y + YF ⎞ v1 = −v o ⎜ L ⎟ ⎝ gm − YF ⎠ Substituting in (1) ⎤ ⎡ ⎛ Y + YF ⎞ ii + v o ⎢ YF + ⎜ L (Yπ + YF ) ⎥ = 0 ⎟ ⎝ gm − YF ⎠ ⎣⎢ ⎦⎥ ∴ B. Murmann vo gm − YF =− ii (gm − YF )YF + (YL + YF )(Yπ + YF ) EE214 Winter 2010-11 – Chapter 7 57 Assuming that gm >> 1/RF = GF gm − YF = gm − (GF + sCF ) ≅ gm − sCF Then gm − sCF vo ≅− ii (gm − sCF )( )(GF + sCF ) + ⎡⎣((GL + GF ) + s(C ( L + CF ) ⎤⎦ ⎡⎣(gπ + GF ) + s(C ( π + CF ) ⎤⎦ Thus vo gm − sC CF ≅− ii a0 + a1s + a 2s2 where a0 = gmGF + (GL + GF )(gπ + GF ) a1 = (gm − GF )CF + (GL + GF )(C π + CF ) + (gπ + GF )(CL + CF ) a 2 = CLC π + CLCF + C π CF B. Murmann EE214 Winter 2010-11 – Chapter 7 58 Further assuming g that gm >> GL, RF >> rπ, Cπ >> CF, and noting g that gm >> gπ a0 ≅ gm RF a1 ≅ gmCF + (GL + GF )C π + gπ (CL + CF ) a 2 ≅ C π (CL + CF ) Then vo R (1− s z1) ≅− F ii 1+ b1s + b 2s2 where ⎛ R + RL ⎞ R τ T + F (CL + CF ) b1 = a1 a0 ≅ RFCF + ⎜ F ⎟ β0 ⎝ RL ⎠ b 2 = a 2 a1 ≅ RF (CL + CF )τ T z1 = + gm CF Often b1 ≅ RFCF B. Murmann EE214 Winter 2010-11 – Chapter 7 59 z1 is a feedforward zero that is located in the RHP. RHP Since it was assumed that Cπ >> CF z1 >> gm ≅ ωT Cπ In these circumstances the feedforward zero can usually be neglected, and the shunt feedback stage has an approximate 2-pole response. vo RF =− ii (1 − s p1 )(1 − s p2 ) The poles p1 and p2 may be complex complex. CF can be used to alter the positions of the poles. B. Murmann EE214 Winter 2010-11 – Chapter 7 60 For example, CF can be chosen to provide a 2-pole MFM response: jω p1 45º σ –B p2 b2 = B. Murmann 1 1 1 = = 2 2 p1p 2 p B 1 EE214 Winter 2010-11 – Chapter 7 61 Thus, ω−3dB = If CF << CL 1 1 = b2 RF (CF + CL )τT ω−3dB ≅ 1 RFCL τT Further insight can be gained from this result by considering the input capacitance of the overall Cherry-Hooper amplifier, which is approximately i t l Cπ/(g /( mRE). ) If we assume that th t CL ≅ k·C k Cπ/(g /( mRE) = kk·τT/RE ⎛ 1 RE ⎞ 1 ⎛ 1 RE ω−3dB ≅ ⎜ =⎜ ⎜ k R ⎟⎟ τ ⎜ kR F ⎠ T F ⎝ ⎝ B. Murmann ⎞ ωT ⎟⎟ ωT = k ⋅ A(0) ⎠ EE214 Winter 2010-11 – Chapter 7 62 Since the overall low-frequency gain, A02, of the cascade is A 02 = RF RE the p per-stage g g gain-bandwidth p product, GB2, is GB2 = (A 02 )1/2 ⋅ ω−3dB ⎛ RF ⎞ ⎛ 1 RE =⎜ ⋅ ⎜ R ⎟⎟ ⎜⎜ k R E ⎠ ⎝ F ⎝ ⎞ ωT ⎟⎟ ωT = k ⎠ Thus, the amplifier approaches the ideal gain-bandwidth product limit for a cascade of identical stages (k=1); there is no bandwidth shrinkage. B. Murmann EE214 Winter 2010-11 – Chapter 7 63 The value of CF that provides a 2-pole MFM response can be found as follows. Realize that ⎛ 1 1⎞ p + p2 b1 = − ⎜ + ⎟ = − 1 p1p 2 ⎝ p1 p 2 ⎠ p1 and p2 are complex conjugates at 45° for an MFM response p1 = σ1 + jω1 p 2 = σ1 − jω1 where −σ1 = ω1 = B. Murmann B 2 EE214 Winter 2010-11 – Chapter 7 64 Thus Th b1 ≅ RFCF = ( 2 )= 2 B B 2 2 B From the earlier analysis, for CF << CL B= 1 1 ≅ b2 RFCL τT Therefore CF = B. Murmann 1 ⎛ 2⎞ ⎜ ⎟= RF ⎝ B ⎠ 2RFCL τT = RF EE214 Winter 2010-11 – Chapter 7 2CL ⋅ τT RF 65 Chapter 8 Noise Analysis y B. Murmann Stanford University Reading Material: Sections 11.1, 11.2, 11.3, 11.4, 11.5, 11.6, 11.7, 11.9 Types of Noise "Man made noise” or interference noise – Signal coupling – Substrate coupling – Finite power supply rejection – Solutions • Fully differential circuits • Layout techniques "Electronic Electronic noise noise" or "device device noise" noise (focus of this discussion) – Fundamental • E.g. "thermal noise" caused by random motion of carriers – Technology related • "Flicker noise" caused by material defects and "roughness" B. Murmann EE214 Winter 2010-11 – Chapter 8 2 Significance of Electronic Noise (1) Signal-to-Noise g Ratio SNR = B. Murmann Psignal Pnoise i ∝ 2 Vsignal 2 Vnoise EE214 Winter 2010-11 – Chapter 8 3 Significance of Electronic Noise (2) Th The "fidelity" "fid lit " off electronic l t i systems t is i often ft determined d t i d by b their th i SNR – Examples • Audio systems • Imagers, Imagers cameras • Wireless and wireline transceivers Electronic noise directly trades with power dissipation and speed Noise has become increasingly important in modern technologies with reduced supply voltages – SNR ~ Vsignal2/Vnoise2 ~ (αVDD)2/Vnoise2 Topics − How to model noise of circuit components − How to calculate/simulate the noise performance of a complete circuit • In which circuits and applications does thermal noise matter? B. Murmann EE214 Winter 2010-11 – Chapter 8 4 Ideal Resistor i(t) () 1V/1kΩ Constant current, independent of time Non-physical – In a physical resistor, carriers "randomly" collide with lattice atoms, giving rise to small current variations over time B. Murmann EE214 Winter 2010-11 – Chapter 8 5 Physical Resistor i(t) 1V/1kΩ "Thermal Noise" or "Johnson Noise" – J.B. Johnson,, "Thermal Agitation g of Electricity y in Conductors," , Phys. y Rev., pp. 97-109, July 1928 Can model random current component using a noise current source in(t) B. Murmann EE214 Winter 2010-11 – Chapter 8 6 Properties of Thermal Noise Present in any conductor Independent p of DC current flow Instantaneous noise value is unpredictable since it is a result of a large number of random, superimposed collisions with relaxation time constants of τ ≅ 0.17ps p – Consequences: • Gaussian amplitude distribution • Knowing g in((t)) does not help pp predict in((t+Δt), ), unless Δt is on the order of 0.17ps (cannot sample signals this fast) • The power generated by thermal noise is spread up to very high frequencies (1/τ ≅ 6,000Grad/s) The only predictable property of thermal noise is its average power! B. Murmann EE214 Winter 2010-11 – Chapter 8 7 Average Power For F a deterministic d t i i ti currentt signal i l with ith period i dT T, th the average power iis T /2 Pav = 1 i2 ( t ) ⋅ R ⋅ dt ∫ T −T /2 This definition can be extended to random signals Assuming a real, stationary and ergodic random process, we can write T /2 1 2 ∫ in ( t ) ⋅ R ⋅ dt T →∞ → T − T /2 Pn = lim For notational convenience, we typically drop R in the above expression and work with "mean mean square" square currents (or voltages) T /2 1 2 ∫ in ( t ) ⋅ dt T →∞ T − T /2 in2 = lim B. Murmann EE214 Winter 2010-11 – Chapter 8 8 Thermal Noise Spectrum Th The so-called ll d power spectral t ld density it (PSD) shows h h how much h power a signal caries at a particular frequency In the case of thermal noise, the power is spread uniformly up to very hi h ffrequencies high i ((about b t 10% d drop att 2 2,000GHz) 000GH ) PSD(f) n0 f The total average noise power Pn in a particular frequency band is found by integrating the PSD f2 Pn = ∫ PSD ( f ) ⋅ df f1 B. Murmann EE214 Winter 2010-11 – Chapter 8 9 Thermal Noise Power Nyquist showed that the noise PSD of a resistor is PSD ( f ) = n0 = 4 ⋅ kT k is the Boltzmann constant and T is the absolute temperature 4kT = 1.66·10-20 Joules at room temperature The total average noise power of a resistor in a certain frequency band is therefore Pn = f2 ∫ 4kT ⋅ df = 4kT ⋅ ( f2 − f1 ) = 4kT ⋅ Δf f1 B. Murmann EE214 Winter 2010-11 – Chapter 8 10 Equivalent Noise Generators We can model the noise using either an equivalent voltage or current generator in2 = v n2 = Pn ⋅ R = 4kT ⋅ R ⋅ Δf Pn 1 = 4kT ⋅ ⋅ Δf R R For R = 1kΩ: For R = 1kΩ: v n2 V2 = 16 ⋅ 10−18 Δf Hz in2 A2 = 16 ⋅ 10−24 Δf Hz v n2 = 4nV / Hz Δf in2 = 4pA / Hz Δf Δf = 1MHz ⇒ v n2 = 4μV B. Murmann Δf = 1MHz ⇒ in2 = 4nA EE214 Winter 2010-11 – Chapter 8 11 Two Resistors in Series ( v n2 = v n1 − v n2 ) 2 2 2 = v n1 + v n2 − 2 ⋅ v n1 ⋅ v n2 Since vn1(t) and vn2(t) are statistically independent, we have 2 2 v n2 = v n1 + vn2 = 4 ⋅ kT ⋅ (R1 + R2 ) ⋅ Δf Al Always remember b to t add dd iindependent d d t noise i sources using i mean squared quantities – Never add RMS values! B. Murmann EE214 Winter 2010-11 – Chapter 8 12 MOSFET Thermal Noise (1) The noise of a MOSFET operating in the triode region is approximately equal to that of a resistor In the saturation region, g , the thermal noise of a MOSFET can be modeled using a drain current source with spectral density i2d = 4kT ⋅ γ ⋅ gm ⋅ Δf For an idealized long channel MOSFET, it can be shown that γ=2/3 For the past 10-15 years, researchers have been debating the value of γ in short channels Preliminary (wrong) results had suggested that in short channels γ can be as high as 5 due to “hot carrier” effects B. Murmann EE214 Winter 2010-11 – Chapter 8 13 MOSFET Thermal Noise (2) [Scholten] At moderate gate bias in strong inversion, short-channel MOSFETS have γ ≅ 1 – A. J. Scholten et al., "Noise modeling for RF CMOS circuit simulation," IEEE Trans. Electron Devices, pp. 618-632, Mar. 2003. – R. P. Jindal, "Compact Noise Models for MOSFETs," IEEE Trans. Electron Devices, e ces, pp pp. 2051-2061, 05 06 , Sep Sep. 2006. 006 B. Murmann EE214 Winter 2010-11 – Chapter 8 14 Thermal Noise in EE214 MOSFET Devices γ ≅ 0.85 γ ≅ 0.7 Parameter γ depends on biasing conditions, but is roughly constant within a reasonable range of gm/ID used for analog design The EE214 HSpice models become inaccurate for sub-threshold operation B. Murmann EE214 Winter 2010-11 – Chapter 8 15 Spice Simulation (1) * EE214 MOS device noise simulation vd d vm vg mn1 h1 dd dd g d c 0 d 0 g 0 0.9 0 9 0 dc 0.7 ac 1 0 0 nmos214 ccvs vm 1 L=0.18u W=10u .op .ac dec 100 10k 1gig .noise v(c) vg .options post brief .inc './ee214_hspice.sp' .end B. Murmann EE214 Winter 2010-11 – Chapter 8 16 Spice Simulation (2) (gm=3.14mS) B. Murmann EE214 Winter 2010-11 – Chapter 8 17 1/f Noise Also called "flicker noise" or "pink noise" – Caused by traps near Si/SiO2 interface that randomly capture and release carriers – Occurs in virtuallyy anyy device,, but is most pronounced p in MOSFETS Several (empirical) expressions exist to model flicker noise – The following expression is used in the EE214 HSpice models 2 i1/ f 2 K f gm Δf = Cox W ⋅ L f For other models, see HSpice manual or – D. Xie et al., "SPICE Models for Flicker Noise in n-MOSFETs from Subthreshold to Strong Inversion," IEEE Trans. CAD, Nov. 2000 Kf is strongly dependent on technology; numbers for EE214: – Kf,NMOS = 0.5·10-25 V2F – Kf,PMOS = 0.25·10-25 V2F B. Murmann EE214 Winter 2010-11 – Chapter 8 18 1/f Noise Corner Frequency B By definition, d fi iti the th frequency f att which hi h the th flicker fli k noise i density d it equals l th the thermal noise density 2 K f gm Δf = 4kTγ ⋅ gm ⋅ Δf Cox W ⋅ L fco ⇒ fco = Kf Kf 1 gm 1 1 ⎛ gm ⎞ ⎛ ID ⎞ = ⎜ ⎟ 4kTγ Cox W ⋅ L 4kTγ Cox L ⎝ ID ⎠ ⎝⎜ W ⎠⎟ For a given gm/ID (e.g. based on linearity considerations), the only way to achieve lower fco is to use longer channel devices − In the above expression, both 1/L and ID/W are reduced for increasing L Example – EE214 NMOS, L = 0.18μm, gm/ID = 12 S/A , ⇒ID/W = 20 A/m ⇒fco = 560 kHz In I newer technologies, t h l i fco can be b on th the order d off 10 MH MHz B. Murmann EE214 Winter 2010-11 – Chapter 8 19 1/f Noise Contribution (1) JJustt as with ith white hit noise, i th the ttotal t l 1/f noise i contribution t ib ti is i found f d by b integrating its power spectral density 2 i1/ f,tot = f2 2 K f gm Δf ∫ Cox W ⋅ L f f 1 = 2 2 ⎛ f ⎞ K gm ⎛f ⎞ K f gm ln ⎜ 2 ⎟ = f 2.3log ⎜ 2 ⎟ Cox W ⋅ L ⎝ f1 ⎠ Cox W ⋅ L ⎝ f1 ⎠ The integrated flicker noise depends on the number of frequency decades – The frequency range from 1Hz …10Hz contains the same amount of flicker noise as 1GHz …10GHz – Note that this is very different from thermal noise So, does flicker noise matter? – Let’s look at the total noise integral (flicker and thermal noise) B. Murmann EE214 Winter 2010-11 – Chapter 8 20 1/f Noise Contribution (2) In the example shown on the left, the noise spectr m is integrated spectrum from 100Hz to 10GHz The contribution of the flicker noise is relatively small, even though its PSD dominates at low frequencies For circuits with very large bandwidth, flicker noise is often insignificant B. Murmann EE214 Winter 2010-11 – Chapter 8 21 Lower Integration Limit Does the flicker noise PSD go to infinity for f → 0? – See e.g. E. Milotti, "1/f noise: a pedagogical review," available at http://arxiv org/abs/physics/0204033 http://arxiv.org/abs/physics/0204033 Even if the PSD goes to infinity, do we care? – Say we are sensing a signal for a very long time (down to a very low frequency), e.g. 1 year ≅ 32 Msec, 1/year ≅ 0.03 μHz – Number of frequency decades in 1/year to 100Hz ≅ 10 – For the example on the previous slide, this means that the integration band changes from 8 to 8+10=18 decades – sqrt(18/8) = 1.5 → Only 50% more flicker noise! B. Murmann EE214 Winter 2010-11 – Chapter 8 22 MOS Model with Noise Generator 2 i2d K f gm 1 = 4kT ⋅ γ ⋅ gm ⋅ + Δf Cox W ⋅ L f B. Murmann EE214 Winter 2010-11 – Chapter 8 Noiseless! (merely a modeling resistor that lets us account for finite dID/dVDS) 23 Other MOSFET Noise Sources G Gate t noise i – "Shot noise" from gate leakage current – Noise due to finite resistance of the gate material – Noise N i due d tto randomly d l changing h i potential/capacitance t ti l/ it b between t th the channel and bulk • Relevant only at very high frequencies • See EE314 Bulk noise Source barrier noise in very short channels – Shot noise from carriers injected across source barrier – R. Navid, C. Jungemann, T. H. Lee and R. W. Dutton, “Highfrequency noise in nanoscale metal oxide semiconductor field effect transistors,” Journal of Applied Physics, Vol. 101(12) , pp. 101-108, June 15, 2007 B. Murmann EE214 Winter 2010-11 – Chapter 8 24 Shot Noise in a PN Junction Shot noise is generally associated with the flow of a DC current In a forward biased diode, shot noise occurs due to randomness in the gy barrier)) carrier transitions across the PN jjunction ((energy The power spectral density of this noise is white up to very high frequencies The noise can be included in the small small-signal signal model as shown below i2 = 2qID ⋅ Δf Constant (“white”) PSD B. Murmann EE214 Winter 2010-11 – Chapter 8 25 Shot Noise in a Bipolar Transistor In a bipolar transistor, the flow of DC current into the base and collector causes shot noise The noise can be modeled via equivalent q current g generators ic2 = 2qIC Δf = 2kTgm Δf g 1 ib2 = 2qIB Δf = 2kT m Δf = 2kT Δf β rπ ic2 ib2 The base and collector noise currents are statistically y independent p as they arise from separate physical mechanisms – This will be important in the context of circuit noise calculations B. Murmann EE214 Winter 2010-11 – Chapter 8 26 BJT Small Signal Model with Noise Generators Thermal noise due to physical rb: Typically negligible Collector shot noise: Base noise components: B. Murmann EE214 Winter 2010-11 – Chapter 8 27 Noise in Circuits (1) Most circuits have more than one relevant noise source In order to quantify the net effect of all noise sources, we must refer the noise sources to a single "interesting" port of the circuit – Usually the output or input In the following discussion, we will first consider only circuits with a perfect voltage drive, i.e. no source resistance RS – Inclusion of finite RS will be discussed later B. Murmann EE214 Winter 2010-11 – Chapter 8 28 Noise in Circuits (2) Output referred noise – Refer noise to output via individual noise transfer functions – Physical concept, exactly what one would measure in the lab B. Murmann Input referred noise – Represent total noise via a fictitious input source that captures all circuit-internal noise sources – Useful for direct comparisons with input signal, or “general purpose” components in which the output noise depends on how the component is used EE214 Winter 2010-11 – Chapter 8 29 Circuit Example For simplicity, let’s neglect – Source impedance – All capacitances it – Burst and flicker noise – ro, rb, and rμ B. Murmann EE214 Winter 2010-11 – Chapter 8 30 Output Referred Noise PSD 1 ⎛ ⎞ 2 v out = ⎜ 4kT Δf + 2qIC Δf ⎟ ⋅ R2 R ⎝ ⎠ ( = 2kTΔf 2R + gmR2 ) ⎛ 2 ⎞ = 2kTgm Δf ⋅ R2 ⎜ + 1⎟ ⎝ gmR ⎠ Shot noise due to base current is absorbed by the input source and does not contribute to noise at the output For large gain (gmR), the collector shot noise dominates B. Murmann EE214 Winter 2010-11 – Chapter 8 31 Input Referred Noise PSD From the previous calculation, we know that Since ⎛ 2 ⎞ 2 = 2kTgm Δf ⋅ R2 ⎜ + 1⎟ v out ⎝ gmR ⎠ v out = A v vin 2 2 v out = A 2v v in where A v = gmR We can write ⎛ 2 ⎞ 2kTgm Δf ⋅ R2 ⎜ + 1⎟ ⎝ gmR ⎠ = 2kT 1 Δf ⎛ 2 + 1⎞ 2 = v in ⎜ ⎟ gm ⎝ gmR ⎠ ( gmR )2 Larger gm translates into lower input referred voltage noise B. Murmann EE214 Winter 2010-11 – Chapter 8 32 Spice Simulation *** EE214 BJT noise example *** biasing ib vcc vb q1 vb vb c1 vb 0 gm=3.67mS, R=10k, Av=36.7 100u 0 npn214 1 *** main circuit v1 vcc 0 vi vi 2.5 vb ac 1 rl vcc vo q2 vo vi 10k 0 npn214 .op .ac dec 100 100 10e9 .noise v(vo) vi .options post brief .inc ‘ee214_hspice.sp' .end d B. Murmann EE214 Winter 2010-11 – Chapter 8 33 Output Referred Noise PSD with Load Capacitance 2 v out 1 1 ⎛ ⎞ = ⎜ 4kT Δf + 2qIC Δf ⎟ ⋅ R || R jωC ⎝ ⎠ 2 1 1 ⎛ ⎞ = ⎜ 4kT Δf + 2qIC Δf ⎟ ⋅ R2 R 1 + jωRC ⎝ ⎠ ⎛ 2 ⎞ 1 = 2kTgm Δf ⋅ R2 ⎜ + 1⎟ ⋅ ⎝ gmR ⎠ 1 + jωRC Same calculation as before before, except that now the noise current drops into parallel combination of R and C Output PSD is shaped by squared magnitude of first order response B. Murmann EE214 Winter 2010-11 – Chapter 8 34 2 2 Input Referred Noise PSD with Load Capacitance S Same calculation l l ti as b before, f exceptt th thatt th the voltage lt gain i iis now frequency dependent 2 vin ( ω) = 2 v out ( ω) A v ( jω ) 2 where A v ( jω ) 2 1 = A v (0) 1 + jωRC ⎛ 2 ⎞ 1 2kTgm Δf ⋅ R2 ⎜ + 1⎟ ⋅ g R 1 + j ω RC ⎝ m ⎠ = 2 1 A 2v ( 0 ) 1 + jωRC 2 2 Input referred noise is frequency independent, because the output noise and gain have the same frequency roll roll-off off 2 ∴ vin = 2kT B. Murmann ⎛ 2 ⎞ 1 Δf ⎜ + 1⎟ gm ⎝ gmR ⎠ EE214 Winter 2010-11 – Chapter 8 35 Spice Simulation *** EE214 BJT noise i example l *** biasing ib vcc vb q1 vb vb c1 vb 0 100u 0 npn214 gm=3.67mS, R=10k, Av=36.7, C=10pF 1 *** main circuit v1 vcc 0 vi vi 2.5 2 5 vb ac 1 rl vcc vo Cl vo 0 q2 vo vi 10k 10p 0 npn214 .op .ac dec 100 100 10e9 .noise v(vo) vi .options post brief .inc 'ee214_hspice.sp' .end B. Murmann EE214 Winter 2010-11 – Chapter 8 36 Signal-to-Noise Ratio Assuming a sinusoidal signal, we can compute the SNR at the output of the circuit using SNR = Psignal Pnoise 1 2 v̂ out v 2 =f 2 2 v ∫ Δoutf ⋅ df f 1 Over which bandwidth should we integrate the noise? Two interesting cases – The output is measured or observed by a system with finite bandwidth (e.g. human ear, or another circuit with finite bandwidth) q y range g of that system y as integration g limits • Use frequency • Applies on a case by case basis – Total integrated noise g noise from zero to “infinite” frequency q y • Integrate B. Murmann EE214 Winter 2010-11 – Chapter 8 37 A Closer Look at The Circuit’s Noise Integral The noise integral converges for upper integration limits that lie beyond the circuit’s pole frequency The total integrated noise (from “0” to “infinity”) is a reasonable metric t i tto use – For convenience in comparing circuits without making bandwidth assumptions – In a circuit where the output is observed without any significant band limiting • E.g. in a sampling circuit • See EE315A,B B. Murmann EE214 Winter 2010-11 – Chapter 8 38 Total Integrated Noise Calculation Let us first consider the noise from the resistor 2 v out,tot 2 ∞ 1 = ∫ 4kTR ⋅ df 1 + j2 π f ⋅ RC 0 ∞ df = 4kTR ∫ 0 1+ = 4kTR ⋅ = ( 2πfRC ) 2 ; du ∫ 1 + u2 = ttan 1 4RC kT C Interesting result – The total integrated noise at the output depends only on C (even g R is g generating g the noise)) though B. Murmann EE214 Winter 2010-11 – Chapter 8 39 Effect of Varying R Increasing R increases the noise power spectral density, but also decreases the bandwidth – R drops out in the end result For C=1pF p ((example p to the right), the total integrated noise is approximately 64μVrms B. Murmann EE214 Winter 2010-11 – Chapter 8 40 −1 u Alternative Derivation The equipartition theorem of statistical mechanics says that each degree of freedom (or energy state) of a system in thermal equilibrium holds an average energy of kT/2 – See e.g. EE248 for a derivation In our circuit, the quadratic degree of freedom is the energy stored on th capacitor the it 1 1 Cv out 2 = kT 2 2 v out 2 = B. Murmann kT C EE214 Winter 2010-11 – Chapter 8 41 Equivalent Noise Bandwidth ⎫ ⎪ ⎪ ⎪ 4kTR ⋅ ΔfENBW ⎬ ⎪ ⎪ 1 = ⎪ 2πRC ⎭ 2 v out,tot = 2 v out,tot f−3dB kT C ΔfENBW = π f−3dB 2 The equivalent noise bandwidth is generally defined as the bandwidth of a brick-wall filter which results in the same total noise power as the filter in question For a simple RC filter, the equivalent noise bandwidth is approximately 1.57 times its 3-dB corner frequency B. Murmann EE214 Winter 2010-11 – Chapter 8 42 Total Integrated Noise Calculation for the Complete Circuit 2 v out,tot = ∞ ∫( ) 4kTR + 2kTgmR2 ⋅ 0 2 1 df 1 + jj2πf ⋅ RC Was 4kTR in previous analysis 2 v out,tot = = kT 4kTR + 2kTgmR2 ⋅ C 4kTR kT ⎛ 1 ⎞ 1 + gmR ⎟ ⎜ C⎝ 2 ⎠ Taking the BJT’s collector shot noise into account, the total integrated noise becomes a multiple of kT/C B. Murmann EE214 Winter 2010-11 – Chapter 8 43 Example SNR Calculation Assumptions – Output carries a sinusoid with 1V peak amplitude – We observe the output without significant band limiting and thus use th total the t t l integrated i t t d noise i in i the th SNR expression i SNR = Psignal Pnoise 1 2 v̂ out 0.5V 2 0.5V 2 2 = = = = 8.59 8 59 ⋅ 106 2 kT ⎛ 1 kT 1 ⎞ ⎛ ⎞ 1+ g R 1 + 3.67mS ⋅ 10kΩ ⎟ ( 763μV ) C ⎜⎝ 2 m ⎟⎠ 10pF ⎜⎝ 2 ⎠ ( ) SNR [ dB] = 10log 8.59 ⋅ 106 = 69.3dB Typical system requirements – Audio: SNR ≅ 100dB – Video: SNR ≅ 60dB – Gigabit Ethernet Transceiver: SNR ≅ 35dB B. Murmann EE214 Winter 2010-11 – Chapter 8 44 Noise/Power Tradeoff Assuming that we're already using the maximum available signal swing, improving the SNR by 6dB means – Increase C by 4x – Decrease R by 4x to maintain bandwidth – Increase gm by 4x to preserve gain – Increase collector current by 4x Bottom line – Improving the SNR in a noise limited circuit by 6dB ("1bit") QUADRUPLES power dissipation ! B. Murmann EE214 Winter 2010-11 – Chapter 8 45 MDS and DR Minimum detectable signal (MDS) – Quantifies the signal level in a circuit that yields SNR=1, SNR 1, i.e. noise power = signal power Dynamic range (DR) is defined as DR = Psignal,max MDS If the noise level in the circuit is independent of the signal level (which is often, but not always the case), it follows that the DR is equal to the " "peak k SNR SNR,"" i.e. i the th SNR with ith the th maximum i signal i l applied li d B. Murmann EE214 Winter 2010-11 – Chapter 8 46 Does Thermal Noise Always Matter? Let’s look at the SNR of an RC circuit with a 1-V sinusoid applied, considering the total integrated noise (kT/C) SNR [dB] 20 40 60 80 100 120 140 C [pF] 0.00000083 0.000083 0.0083 0 83 0.83 83 8300 830000 Hard to make such small capacitors… Designer will be concerned about thermal noise; i componentt sizes i often ft sett by b SNR A difficult battle with thermal noise … Rules of thumb – Up to SNR ~ 30-40dB, integrated circuits are usually not limited by thermal noise – Achieving SNR >100dB is extremely difficult • Must usually rely on external components, or reduce bandwidth and remove noise by a succeeding filter • See S e.g. oversampling li ADC ADCs iin EE315B B. Murmann EE214 Winter 2010-11 – Chapter 8 47 More on Input Referred Noise Suppose you wanted to sell this amplifier as a “general purpose” building block R How would you communicate information about its noise performance to the customer? None of the metrics that we have used so far will work to describe the circuit independent of the target application – The computed input and output referred voltage noise assumed that the circuit is driven by an ideal voltage source B. Murmann EE214 Winter 2010-11 – Chapter 8 48 Two-Port Representation Using Equivalent Voltage and Current Noise Generators Short Short-circuit circuit both inputs and equate output noise – This yields vi2 Open-circuit both inputs and equate output noise – This yields ii2 This representation is valid for “any” source impedance Sometimes need to consider correlation between equivalent voltage and current generator, but often times only one of the two generators matters in the target application If both generators matter (and they are correlated), it is usually best to avoid working with input referred noise representations B. Murmann EE214 Winter 2010-11 – Chapter 8 49 Datasheet Example B. Murmann EE214 Winter 2010-11 – Chapter 8 50 Examples Single BJT device Single g MOS device CE stage CB stage B. Murmann EE214 Winter 2010-11 – Chapter 8 51 BJT Input Voltage Noise (1) To find input referred voltage generator generator, short the input of both circuit models and equate output noise Neglecting Cμ, rc and re for simplicity Text, p. 757 ( 2 2 2 2 io1 ≅ ic2 + gm ib ⋅ rb + v b2 ) 2 2 2 io2 ≅ gm vi 2 2 io1 = io2 B. Murmann ⇒ v i2 ≅ v b2 + ic2 2 gm EE214 Winter 2010-11 – Chapter 8 + ib2 ⋅ rb2 52 BJT Input Voltage Noise (2) The PSD of the BJT input voltage noise generator is therefore 2qI vi2 ≅ 4kTrb + 2C + 2qIBrb2 Δf gm ≅ 4kTrb + ≅ 4kTr 4kT b + 2 2⎤ 2qIC ⎡ gm rb 1 + ⎢ ⎥ 2 β ⎥⎦ gm ⎢⎣ qC 2qI 2 gm ⎛ 1 ⎞ ≅ 4kT ⎜ rb + ⎟ 2gm ⎠ ⎝ B. Murmann EE214 Winter 2010-11 – Chapter 8 53 BJT Input Current Noise (1) To find input referred current generator, open circuit the input of both circuit models and equate output noise Text, p. 757 2 2 2 io1 = ic2 + gm ⋅ ib ⋅ z π 2 2 2 io2 = gm ⋅ ii ⋅ z π 2 2 = io2 io1 B. Murmann ⇒ ii2 = ib2 + 2 2 ic2 2 gm zπ EE214 Winter 2010-11 – Chapter 8 2 54 BJT Input Current Noise (2) ii2 Δf = 2qIB + = 2qIB + 2qIC 2 gm zπ 1 β ( jω ) jωCπ rπ = = where z π = 1 1 + jωrπCπ gm rπ + jωCπ rπ ⋅ 2 2qIC β ( jω ) 2 2 ⎛ ⎛ ⎞ ⎞ ⎜ 1+ ⎜ ω ⎟ ⎟ ⎜ ωβ ⎟ ⎟ ⎜ ⎝ ⎠ ⎟ = 2qIB ⎜ 1 + β0 ⎜ ⎟ ⎜ ⎟ ⎜ ⎟ ⎝ ⎠ where ωβ = ω 1 ≅− T rπCπ β0 The term due to IC is negligible at low frequencies frequencies, but becomes comparable to the base current contribution at ωb = ωβ β0 ≅ B. Murmann ωT β0 EE214 Winter 2010-11 – Chapter 8 55 Plot of BJT Input Noise Current PSD Text, p. 760 N l t (f Neglect (for BJT) B. Murmann EE214 Winter 2010-11 – Chapter 8 56 MOS Input Voltage Noise 2 io1 = ic2 Text, p. 762 2 2 2 io2 = gm vi v i2 = B. Murmann ic2 2 gm v i2 Kf 1 1 = 4kTγ + Δf gm WLCox f EE214 Winter 2010-11 – Chapter 8 57 MOS Input Current Noise 2 io1 = ic2 2 + gm ⋅ i2g 2 io2 ii2 B. Murmann = i2g ⎛ ω +⎜ ⎜ ⎝ ωT 2 ⎞ 2 ⎟⎟ ic ⎠ ⎛ f ii2 ≅ 2qIG + ⎜ ⎜ Δf ⎝ fT ⎞ ⎟⎟ ⎠ 2 EE214 Winter 2010-11 – Chapter 8 ⋅ 1 ω2C2gs ≅ ii2 ≅ ic2 ⎛ω ⎞ ⋅⎜ T ⎟ ⎝ ω ⎠ 2 ⎛ K f gm 4kT g γ + ⎜⎜ m WLCox ⎝ + i2g ⎛ω ⎞ ⋅⎜ T ⎟ ⎝ ω ⎠ 2 1⎞ ⎟ f ⎠⎟ 58 2 BJT versus MOS (1) Noiseless Transistor Consider low source impedance Æ voltage noise will dominate ⎡ v2 ⎤ ⎛ 1 ⎞ ⎢ i ⎥ ≅ 4kT ⎜ rb + ⎟ 2gm ⎠ ⎢⎣ Δf ⎥⎦ ⎝ BJT ⎡ v2 ⎤ Kf 1 1 ⎢ i ⎥ ≅ 4kTγ + gm WLCox f ⎢⎣ Δf ⎦⎥ MOS BJT is usually y superior p – Need less gm for approximately same noise – gm/I is higher, making it easier to achieve low noise at a given current budget B. Murmann EE214 Winter 2010-11 – Chapter 8 59 BJT versus MOS (2) Consider high source impedance Æ current noise will dominate ⎛ ⎞ ⎡ i2 ⎤ IC ⎟ ⎢ i ⎥ ≅ 2q ⎜ IB + 2 ⎜ Δf β ( jω) ⎠⎟ ⎣⎢ ⎦⎥BJT ⎝ 2 ⎡ i2 ⎤ ⎛ f ⎞ ⎛ K g2 i ⎢ ⎥ ≅ 2qIG + ⎜ ⎟ ⎜ 4kTγgm + f m ⎜ WLCox Δf ⎝ fT ⎠ ⎝ ⎣⎢ ⎦⎥MOS MOS is usually superior – Gate leakage current (IG) is typically much smaller than BJT base current thin, as e e.g. g in a 45-nm 45 nm • Unless the gate oxide becomes very thin CMOS process that does not use high-k gate dielectrics B. Murmann EE214 Winter 2010-11 – Chapter 8 60 1⎞ ⎟ f ⎟⎠ Example Revisited (With finite RS) ⎛ vi2 1 1 ⎞ ≅ 4kT ⎜ rb + + 2 ⎟ ⎜ Δf 2gm gmR ⎟⎠ ⎝ B. Murmann 1 ⎛ ⎞ 4kT I ii2 C R ⎟+ = 2q ⎜ IB + 2⎟ 2 ⎜ Δf β ( jω ) ⎠ β ( j ω ) ⎝ EE214 Winter 2010-11 – Chapter 8 61 Calculation of Equivalent Source Noise 2 v 2s = vRs + v i2 + R2s ii2 1 ⎡ ⎛ ⎞ 4kT ⎤⎥ ⎛ v 2s I 1 1 ⎞ 2⎢ C R ⎟+ = 4kTRS + 4kT ⎜ rb + + 2 ⎟ + Rs ⎢2q ⎜ IB + 2⎟ 2⎥ ⎜ ⎟ ⎜ 2g Δf g R m β ( jω ) ⎠ β ( jω ) ⎥ ⎢ ⎝ m ⎠ ⎝ ⎣ ⎦ B. Murmann EE214 Winter 2010-11 – Chapter 8 62 Alternative Method Draw complete circuit with all noise sources Compute transfer function from each noise source to the output Refer to output using squared transfer function Sum all output referred noise components Divided obtained sum by squared transfer function from input source to the output B. Murmann EE214 Winter 2010-11 – Chapter 8 63 Output Referred Noise PSD H(s) = v out zπ = −gmR vs z π + Rs =− 2 v out 1 2 ⎛ ⎞ = 4kTRS* H(s) + ⎜ 4kT + 2qIC ⎟ ⋅ R2 R Δf ⎝ ⎠ gmR g R 1+ m s β ( jω ) (neglecting base shot noise for simplicity) 2 1 ⎛ ⎞ 2 v out ⎜ 4kT R + 2qIC ⎟ ⋅ R ⎠ = Δf 2 = 4kTRS* + ⎝ 2 Δf H(s) H(s) v 2s B. Murmann EE214 Winter 2010-11 – Chapter 8 64 Source Referred Noise PSD v 2s = 4kTRS* + Δf gmRs 1 ⎛ ⎞ 2 ⎜ 4kT R + 2qIC ⎟ ⋅ R 1 + β jω ( ) ⎝ ⎠ 2 ( gmR )2 2⎞ ⎛ ⎡ 1 gmRs ⎟ 1 ⎤ ⎜ = 4kTRS + 4kT rb + ⎢ + 2 ⎥ ⋅ 1+ ⎜ β ( jω ) ⎟ ⎢⎣ 2gm gmR ⎥⎦ ⎝ ⎠ This result does not match what we would expect from the analysis with BJT input referred generators: 1 ⎤ ⎡ 4kT ⎥ ⎛ ⎢ v 2s 2qI 1 1 ⎞ C R = 4kTRS + 4kT ⎜ rb + + 2 ⎟ + Rs2 ⎢ + 2 2⎥ ⎜ ⎟ Δf 2g gmR ⎠ m β ( jω ) ⎥ ⎢ β ( jω ) ⎝ ⎣ ⎦ B. Murmann EE214 Winter 2010-11 – Chapter 8 65 Reason for Discrepancy It turns t outt that th t the th resultlt obtained bt i d using i the th BJT input i t referred f d generators is not quite correct From vi2 From ii2 1 ⎤ ⎡ 4kT ⎥ ⎛ v 2s 1 1 ⎞ 2 ⎢ 2qIC R = 4kTRS + 4kT ⎜ rb + + 2 ⎟ + Rs ⎢ + 2 2⎥ ⎜ ⎟ Δf 2 2g g R m β ( jω ) ⎥ ⎢ β ( jω ) m ⎠ ⎝ ⎣ ⎦ Due to RS Due to rb Collector shot noise Due to R Collector shot noise Due to R In this expression, expression we are adding the powers of correlated noise currents without taking the correlation into account! This leads to an expression that overestimates the noise B. Murmann EE214 Winter 2010-11 – Chapter 8 66 Conclusion on Input Referred Noise Generators W Working ki with ith input i t referred f d noise i generators t saves time ti and d works k wellll if the input source can be modeled close to a an ideal voltage source (RS is small) or an ideal current source (RS is large) – In this case, case either the input referred noise voltage or current will clearly dominate and only one of the two generators must be considered – Note that for RS=0, the two expression on slide 65 match perfectly For any scenario in-between, working with input referred generators can still work as long as the input referred current and voltage generators are statistically independent, i.e. they are due physically distinct noise mechanisms – This is not the case for the expressions of slide 65 When all of these conditions fail, it is a must to analyze y the circuit from first principles, i.e. consider all noise sources at their root and refer them to the point(s) of interest via their individual transfer functions B. Murmann EE214 Winter 2010-11 – Chapter 8 67 Noise Performance of Feedback Circuits: Ideal Feedback Consider an amplifier with input referred noise sources placed into an ideal series-shunt feedback f configuration f (no ( loading effects) ff ) We can identify the equivalent input noise generators for the overall circuit in the same way we have done this previously – Short-circuit the input, find input noise voltage generator that yields the same output noise in both circuits – Open-circuit the input, find input noise current generator that yields the same output noise in both circuits B. Murmann EE214 Winter 2010-11 – Chapter 8 68 Input Shorted Equal output noise in both circuits is obtained for 2 vi2 = via B. Murmann EE214 Winter 2010-11 – Chapter 8 69 Input Open Equal output noise in both circuits is obtained for 2 ii2 = iia B. Murmann EE214 Winter 2010-11 – Chapter 8 70 Observation For the idealized example studied on the previous slides, we see that the equivalent input noise generators of the amplifier can be moved unchanged h d outside t id th the ffeedback db k lloop – Applying feedback has no effect on the circuit’s noise performance – Note that this is very different from the effect of feedback on distortion performance This results holds for all four possible (ideal) feedback configurations – Prove this as an exercise In practical feedback configurations, the input referred generators must be computed while taking loading effects into account – Loading makes the calculations more complicated, and generally worsens the th noise i performance f – In the best possible design outcomes, the noise performance can approach (but not surpass) that of an idealized configuration B. Murmann EE214 Winter 2010-11 – Chapter 8 71 Including Loading and Noise from the Feedback Network Th The mostt generall way off including i l di loading l di and d noise i ffrom th the ffeedback db k network is to apply the same procedure as before – Short-circuit the input, find input noise voltage generator that yields the same output noise in both circuits – Open-circuit the input, find input noise current generator that yields the same output noise in both circuits A more convenient way to include loading and noise from the feedback network is to use the same two-port approximations we have already utilized for transfer function analysis Procedure – Absorb loading effects into the basic amplifier and work with ideal feedback network – Re Re-compute compute the input referred noise generators of the basic amplifier in presence of loading – Using the previous result, the re-computed noise generators of the basic amplifier can now be moved outside the feedback loop B. Murmann EE214 Winter 2010-11 – Chapter 8 72 Example: Practical Series-Shunt Circuit B. Murmann EE214 Winter 2010-11 – Chapter 8 73 Basic Amplifier with Loading Step 1 – Redraw the basic amplifier with loading included B. Murmann EE214 Winter 2010-11 – Chapter 8 74 Input Referred Noise Current (Open-Circuited Input) Step 2 – Compute input referred current noise – The result corresponds to the desired ii 2 ii2 = iia B. Murmann EE214 Winter 2010-11 – Chapter 8 75 Input Referred Noise Voltage (Short-Circuited Input) Step 3 – Compute input referred voltage noise – The result corresponds to the desired vi via2 iia2 RE||RF 2 2 2 vi2 = v ia + iia ⋅ (RE || RF ) + 4kT (RE || RF ) Δf 4kT(RE||RF) B. Murmann EE214 Winter 2010-11 – Chapter 8 76 Example: Practical Shunt-Shunt Circuit 2 v i2 = via ii2 2 = iia + 2 via RF2 + 4kT 1 Δf RF (assuming iia and via are uncorrelated) B. Murmann EE214 Winter 2010-11 – Chapter 8 77 High Frequency Issue At low l ffrequencies, i it iis ttypically i ll nott h hard d tto minimize i i i th the iinputt noise i current contribution due to via However, at high frequencies, any shunt capacitance at the input tends t make to k the th via contribution t ib ti more significant i ifi t Means that via must be minimized in high speed circuits, typically by increasing gm of the input device Æ higher power dissipation 2 ii2 B. Murmann 2 = iia 2 + v ia 1 1 + jωCi + 4kT Δf RF RF EE214 Winter 2010-11 – Chapter 8 (assuming iia and via are uncorrelated) 78 Local Feedback Circuit Examples Av = vo g R = m vi 1 + gmR Gm = io gm = vi 1 + gmR Ai = io g R = m ii 1 + gmR Neglecting finite ro, backgate effect and flicker noise for simplicity B. Murmann EE214 Winter 2010-11 – Chapter 8 79 Source Follower Th The noise i performance f can be b analyzed l d by b treating t ti the th circuit i it as a series feedback stage (see text, sections 8.6.2 and 11.7.2) – We will do a direct analysis instead 2 ( v o2 = i2d + ir2 ) ⎛ ⎞ ⎜ 1 ⎟ i2d + ir2 2 ⋅⎜ ⎟ = 2 ⋅ Av gm ⎜ gm + 1 ⎟ R⎠ ⎝ Often negligible v i2 = i2d 2 gm + ir2 2 gm = 4kT ⎛ 1 1 ⎞ Δf ⎜ γ + ⎟ gm ⎝ gmR ⎠ The noise in a resistively loaded source follower is typically dominated by the contribution from the transistor The input referred noise voltage can be approximated by the drain current noise reflected through the device’s device s transconductance B. Murmann EE214 Winter 2010-11 – Chapter 8 80 Degenerated Common Source Stage io = id + ( ir − id ) = ir ⎛ i2 ⎞ 2 = ⎜ d2 + ir2R2 ⎟ ⋅ Gm ⎜ gm ⎟ ⎝ ⎠ v i2 = i2d 2 gm + ir2R2 = 4kTγ gm + 1 R gmR 1 + id 1 + gmR 1 + gmR = ir GmR + id io2 gm Gm gm 1 Δf + 4kTRΔf gm The input referred voltage noise consists of drain current noise, reflected through gm, plus the resistor’s voltage noise B. Murmann EE214 Winter 2010-11 – Chapter 8 81 Common Gate Stage io = ir gmR 1 + id 1 + gmR 1 + gmR = ir A i + id Ai gmR Often negligible io2 = ⎛ i2 ⎜ d 2 2 ⎜ gm ⎝ R ⎞ + ir2 ⎟ ⋅ A i2 ⎟ ⎠ ii2 = i2d 2 2 gm R + ir2 = 4kT 1 ⎛ γ ⎞ Δf ⎜ 1 + ⎟ R ⎝ gmR ⎠ Th The input i t referred f d currentt noise i from f the th transistor t i t is i often ft negligible li ibl (at ( t low frequencies) The noise tends to be dominated by the devices providing the source and dd drain i bi bias currents t ((resistors i t or currentt sources)) B. Murmann EE214 Winter 2010-11 – Chapter 8 82 Common Gate Stage at High Frequencies 1 + j ωC 1 R 2 ii = 4kT Δf + 4kTγgm Δf 2 R gm ⎛ ωC ⎞ 1 ≅ 4kT Δf + 4kTγgm Δf ⎜ ⎟ R ⎝ gm ⎠ 2 2 The input referred current noise from the transistor can be significant at high frequencies (near the cutoff frequency of the current transfer) B. Murmann EE214 Winter 2010-11 – Chapter 8 83 Summary – Noise in Feedback Circuits Applying ideal feedback around an amplifier does not alter its input referred noise performance In practical circuits, loading and noise from the feedback network tend to deteriorate the circuit’s overall noise performance – This is especially true at high frequencies, where parasitic capacitances it can increase i th the noise i ttransfer f from f sources that th t are typically negligible at low frequencies Loading effects can be considered by applying the same two-port approximation i ti methods th d used d iin th the ttransfer f ffunction ti analysis l i off practical ti l feedback amplifiers – Absorb loading and feedback network noise sources into forward amplifier and work with idealized feedback result B. Murmann EE214 Winter 2010-11 – Chapter 8 84 Additional Topics in Noise Analysis Covered in EE314 – RF-centric metrics • Noise figure • Receiver sensitivity – Phase noise in oscillators Covered in EE315A,B – Noise in filters and switched capacitor circuits Other – Cyclostationary noise • Noise in circuits that are driven by a periodic waveform that modulates the power spectral densities • E.g. mixers B. Murmann EE214 Winter 2010-11 – Chapter 8 85 Chapter 9 Distortion Analysis y B. Murmann Stanford University Reading Material: Sections 1.4.1, 5.3.2 Overview Low frequency distortion analysis Effect of feedback on distortion High frequency distortion analysis B. Murmann EE214 Winter 2010-11 – Chapter 9 2 Introduction All electronic l t i circuits i it exhibit hibit some llevell off nonlinear li b behavior h i – The resulting waveform distortion is not captured in small-signal models In the first section of this chapter, we will begin by looking at the basic tools needed to analyze “memoryless” nonlinearities, i.e. nonlinearities that can be represented by a frequency independent model – Such models are valid in a frequency range where all capacitances and inductances in the circuit of interest can be ignored As a driving example, we will analyze the nonlinearity in the V-I transduction of BJTs and MOSFETs The general approach taken is to model the nonlinearities via a power series that links the input and output of the circuit – This approach is useful and accurate for the case of “small small distortion” distortion and cannot be used to predict the effect of gross distortion, e.g. due to signal clipping B. Murmann EE214 Winter 2010-11 – Chapter 9 3 Small-Signal AC Model io Io = IOQ + io + Vi = VIQ + vi - gm = dIo dVi V = V i IQ = f '(VIQ ) Io = f(Vi) gm gm⋅vi + vi - io gm vi VIQ B. Murmann Vi EE214 Winter 2010-11 – Chapter 9 4 Taylor Series Model (3) f '(VIQ ) f ''(VIQ ) 2 f (VIQ ) f(Vi ) = f(VIQ ) + (Vi − VIQ ) + (Vi − VIQ ) + (Vi − VIQ )3 + ... 1! 2! 3! f2(Vi) f3(Vi) f(Vi) f2(Vi) f3(Vi) B. Murmann Vi VIQ EE214 Winter 2010-11 – Chapter 9 5 Relationship Between Incremental Variables Using vi = Vi − VIQ io = Io − IOQ = f(Vi ) − f(VIQ ) and we obtain io = a1v i + a2 v i2 + a3 v i3 + ... where f (m) (VIQ Q) am = m! Note that a1 ≡ gm a2 ≡ 1 ' gm 2 a3 ≡ 1 '' gm 6 In practice, it is often sufficient to work with a truncated nth order power series io ≅ a1v i + a2 v i2 + ... + an v in B. Murmann EE214 Winter 2010-11 – Chapter 9 6 Graphical Illustration io n=2 vi n→∞ n=3 A model that relates the incremental signal components (vi, io) though a nonlinear expression is sometimes called “large-signal AC model” The accuracy of a truncated power series model depends on the signal range and the curvature of the actual transfer function – Using a higher order series generally helps, but also makes the analysis more complex – As we will see, using a third order series is often sufficient to model th relevant the l t distortion di t ti effects ff t in i practical, ti l weakly kl nonlinear li circuits i it B. Murmann EE214 Winter 2010-11 – Chapter 9 7 Harmonic Distortion Analysis Apply a sinusoidal signal and collect harmonic terms in the output signal v i = vˆ i ⋅ cos ( ωt ) 2 3 io = a1vˆ i cos ( ωt ) + a2 ⎡⎣ vˆ i cos ( ωt ) ⎤⎦ + a3 ⎡⎣ vˆ i cos ( ωt ) ⎤⎦ + ... cos2 ( α ) = 1 ⎡cos ( 2α ) + 1⎤⎦ 2⎣ cos3 ( α ) = 1 ⎡cos ( 3α ) + 3 cos ( α ) ⎤⎦ 4⎣ ⎡1 ⎤ ∴ io = ⎢ a2 vˆ i2 ⎥ ⎣2 ⎦ B. Murmann DC shift 3 ⎡ ⎤ + ⎢a1vˆ i + a3 vˆ i3 ⎥ cos ( ωt ) 4 ⎣ ⎦ Fundamental ⎡1 ⎤ ⎡1 ⎤ + ⎢ a2 vˆ i2 ⎥ cos ( 2ωt ) + ⎢ a3 vˆ i3 ⎥ cos ( 3ωt ) + ... ⎣2 ⎦ ⎣4 ⎦ Harmonics EE214 Winter 2010-11 – Chapter 9 8 Observations Th The quadratic d ti tterm ((a2) give i rises i tto an undesired d i d second dh harmonic i ttone and a DC shift The cubic term (a3) give rises to an undesired third harmonic tone and it also l modifies difi th the amplitude lit d off th the ffundamental d t l – a3 < 0 Æ “gain compression” – a3 > 0 Æ “gain expansion” y x + 0.3x3 x x - 0.3x 0 3x3 x B. Murmann EE214 Winter 2010-11 – Chapter 9 9 Waveforms with Gain Expansion x = cos ( 2πt ) B. Murmann a1 = 1 a3 = 0.3 EE214 Winter 2010-11 – Chapter 9 10 Waveforms with Gain Compression x = cos ( 2πt ) B. Murmann a1 = 1 a3 = −0.3 EE214 Winter 2010-11 – Chapter 9 11 Higher Order Terms cosm ( α ) = 1 m 2 ( e jα + e − j α ) m = 1 m 2 m ⎛m⎞ ∑ ⎜ k ⎟e jkα e k =0 ⎝ − j( m − k ) α ⎠ cos4 ( α ) = 1 ⎡cos ( 4α ) + 4 cos ( 2α ) + 3 ⎤⎦ 8⎣ cos5 ( α ) = 1 ⎡cos ( 5α ) + 5cos ( 3α ) + 10 cos ( α ) ⎤⎦ 16 ⎣ Can show that – Terms raised to even powers of m affect the DC shift and even harmonics up to m – Terms raised to odd powers of m affect the fundamental and odd harmonics up to m B. Murmann EE214 Winter 2010-11 – Chapter 9 12 Inspection of 4th and 5th Order Contributions 3 ⎡1 ⎤ ∴ io = ⎢ a2 vˆ i2 + a4 vˆ i4 ⎥ 8 ⎣2 ⎦ As long as 3 5 ⎡ ⎤ a5 vˆ i5 ⎥ cos ( ωt ) + ⎢a1vˆ i + a3 vˆ i3 + 4 16 ⎣ ⎦ 1 ⎡1 ⎤ + ⎢ a2 vˆ i2 + a4 vˆ i4 ⎥ cos ( 2ωt ) 2 2 ⎣ ⎦ 5 ⎡1 ⎤ + ⎢ a3 vˆ i3 + a5 vˆ i5 ⎥ cos ( 3ωt ) 16 ⎣4 ⎦ ⎡1 ⎤ + ⎢ a 4 vˆ i4 ⎥ cos ( 4ωt ) ⎣8 ⎦ ⎡1 ⎤ + ⎢ a5 vˆ i5 ⎥ cos ( 5ωt ) ⎣ 16 ⎦ B. Murmann a 4 vˆ i4 << a2 vˆ i2 and a5 vˆ i5 << a3 vˆ i3 or equivalently vˆ i << a2 a4 and vˆ i << a3 a5 the 4th and 5th order terms can be neglected This condition is usually met in practical, weakly nonlinear circuits EE214 Winter 2010-11 – Chapter 9 13 Fractional Harmonic Distortion Metrics HD2 = amplitude of second harmonic distortion signal amplitude of fundamental HD3 = amplitude amplit de of third harmonic distortion signal amplitude of fundamental IIncluding l di only l contributions t ib ti ffrom tterms up tto 3rdd order, d th these quantities titi become 1 a2 vˆ i2 1 a2 2 HD2 ≅ ≅ vˆ i 3 2 a 3 1 ˆ ˆ a1v i + a3 vi 4 1 a3 vˆ i3 1 a3 2 4 ≅ HD3 ≅ vˆ i 3 4 a 3 1 a1vˆ i + a3 vˆ i 4 B. Murmann EE214 Winter 2010-11 – Chapter 9 14 Total Harmonic Distortion THD = total power of distortion signals power of fundamental = HD22 + HD32 + HD24 + ... THD iis often ft d dominated i t db by th the HD2 and/or d/ HD3 term t Typical application requirements – Telephone audio: THD < ~10% – Video: THD < ~1% – RF low noise amplifiers: THD < ~0.1% – High quality audio: THD < ~0.01% B. Murmann EE214 Winter 2010-11 – Chapter 9 15 Intermodulation Distortion (1) Consider applying two tones to the nonlinear device v i = vˆ i1 ⋅ cos ( ω1t ) + vˆ i2 ⋅ cos ( ω2 t ) io = a1 ⎡⎣ vˆ i1 ⋅ cos ( ω1t ) + vˆ i2 ⋅ cos ( ω2 t ) ⎤⎦ +a2 ⎣⎡ vˆ i1 ⋅ cos ( ω1t ) + vˆ i2 ⋅ cos ( ω2 t ) ⎦⎤ 2 3 +a3 ⎡⎣ vˆ i1 ⋅ cos ( ω1t ) + vˆ i2 ⋅ cos ( ω2 t ) ⎤⎦ + ... Inspect second-order term 2 a2 ⎡⎣ vˆ i1 ⋅ cos ( ω1t ) + vˆ i2 ⋅ cos ( ω2 t ) ⎤⎦ = a2 ⎡⎣ vˆ i1 ⋅ cos ( ω1t ) ⎤⎦ 2 +a2 ⎡⎣ vˆ i2 ⋅ cos ( ω2 t ) ⎤⎦ 2 C Causes HD +2a2 ⎡⎣ vˆ i1vˆ i2 ⋅ cos ( ω1t ) cos ( ω2 t ) ⎤⎦ B. Murmann EE214 Winter 2010-11 – Chapter 9 New 16 Second-Order Intermodulation 2a2 ⎡⎣ vˆ i1vˆ i2 ⋅ cos ( ω1t ) cos ( ω2 t ) ⎤⎦ = a2 vˆ i1vˆ i2 ⎡⎣cos ({ω1 + ω2 } t ) + cos ({ω1 − ω2 } t ) ⎤⎦ The output will contain tones at the sums and differences of the applied frequencies We define the fractional second-order intermodulation as IM2 = = amplitude of second second-order order IM components (for vˆ i1 = vˆ i2 = vˆ i ) amplitude of fundamentals a2 vˆ i2 a2 = vv̂ i a1 vˆ i a1 = 2HD2 B. Murmann EE214 Winter 2010-11 – Chapter 9 17 Third-Order Intermodulation (1) 3 a3 ⎡⎣ vˆ i1 ⋅ cos ( ω1t ) + vˆ i2 ⋅ cos ( ω2 t ) ⎤⎦ = a3 ⎡⎣ vˆ i1 ⋅ cos ( ω1t ) ⎤⎦ 3 +a3 ⎡⎣ vˆ i2 ⋅ cos ( ω2 t ) ⎤⎦ 3 Causes HD 2 +3a3 ⎡ vˆ i1vˆ i2 ⋅ cos ( ω1t ) cos2 ( ω2 t ) ⎤ ⎣ ⎦ 2ˆ +3a3 ⎡ vˆ i1 v i2 ⎣ ⋅ cos 2 New ( ω1t ) cos ( ω2t )⎤⎦ 2 3 3 ⎡ vˆ i1vˆ i2 3a ⋅ cos ( ω1t ) cos2 ( ω2 t ) ⎤ ⎣ ⎦ = 3 2 ⎡ ⎤ a3 vˆ i1vˆ i2 ⎣2cos ( ω1t ) + cos ({2ω2 − ω1} t ) + cos ({2ω2 + ω1} t ) ⎦ 4 “Gain desensitization term” For a3<0, large vi2 reduces fundamental tone due to vi1 B. Murmann Third-order Intermodulation Products EE214 Winter 2010-11 – Chapter 9 18 Third-Order Intermodulation (2) Third-order intermodulation products appear at (2ω2 ± ω1) and (2ω1 ± ω2) We define the fractional third-order intermodulation as IM3 = = amplitude of third-order IM components (for vˆ i1 = vˆ i2 = vˆ i ) amplitude of fundamentals 3 a3 vˆ i3 3 a3 2 = v̂ i = 3HD3 4 a1 vˆ i 4 a1 Note that for ω2 ≅ ω1, the third-order intermodulation products are close to the original frequencies and cannot be filtered out – This is a significant issue in narrowband systems B. Murmann EE214 Winter 2010-11 – Chapter 9 19 Distortion in a CE Stage (1) Ic = Ise Vbe VT kT VT = q dIc a1 = dVbe = a2 = am = B. Murmann Vbe Vbe = VBEQ I = s e VT VT Vbe = VBEQ ICQ ≡ gm VT 1 d2Ic 2 2 dVbe = Vbe = VBEQ 1 ICQ 2 VT2 1 ICQ m! VTm EE214 Winter 2010-11 – Chapter 9 20 Distortion in a CE Stage (2) 1 a3 2 1 ⎛ vˆ be ⎞ HD3 ≅ vˆ be = ⎜ ⎟ 4 a1 24 ⎜⎝ VT ⎟⎠ 1 a2 1 vˆ be HD2 ≅ vˆ be = 2 a1 4 VT 2 Low distortion in the collector current quires the B-E voltage excursion to be much smaller than VT ≅ 26mV Checking for the valid range of a third order model yields vˆ be << a2 = 12VT a4 and vˆ be << a3 = 20VT a5 For a B-E voltage swing of VT, we have HD2 ≅ 25% HD3 ≅ 4.17% Note that a typical application will demand much lower distortion B. Murmann EE214 Winter 2010-11 – Chapter 9 21 Distortion in a CS Stage (1) Id = 1 W μCox Vgs − Vt 2 L ( ) 2 a1 = dId dVgs = μCox a2 = = μCox Vgs = VGSQ W Vgs − Vt L ( ) Vgs = VGSQ 2I W VOV = DQ ≡ gm L VOV 1 d2Id 2 2 dVgs = Vgs = VGSQ 1 W IDQ μCox = 2 2 L VOV a3 = 0 B. Murmann EE214 Winter 2010-11 – Chapter 9 22 Distortion in a CS Stage (2) HD2 ≅ 1 a2 1 v̂ gs vˆ gs = 2 a1 4 VOV HD3 = 0 Small second harmonic distortion in the drain current requires the G-S voltage excursion to be much smaller than the quiescent point gate overdrive d i (VOV=V VGS-V Vt) An idealized square-law device does not introduce high order distortion – However, this is not true for a real short-channel MOSFET Relevant effects – Velocity saturation, mobility reduction due to vertical field – Biasing g in moderate or weak inversion – Nonlinearity in the device’s output conductance – … B. Murmann EE214 Winter 2010-11 – Chapter 9 23 Distortion in Modern MOSFETS Distortion in modern MOSFET devices is generally hard to model accurately A basic approach for devices operating in strong inversion is to include short channel effects via basic extensions to the square law model – E.g. model velocity saturation as resistive source degeneration – See e e.g. g Terrovitis & Meyer, Meyer JSCC 10/2000 Another approach is to extract the coefficients from “known-to-beaccurate” Spice models – Find coefficients am by simulating the derivatives of I-V curves – See e.g. Blaakmeer et al., JSSC 6/2008 Unfortunately, generating accurate Spice models for MOSFET distortion is a very difficult task – See e.g. R. van Langevelde et al., IEDM 2000 Never trust a Spice model blindly! B. Murmann EE214 Winter 2010-11 – Chapter 9 24 Distortion of EE214 NMOS Device B. Murmann EE214 Winter 2010-11 – Chapter 9 25 Distortion in a BJT Differential Pair (1) ⎛ V ⎞ Icd = Ic1 − Ic2 = αIEE tanh ⎜ id ⎟ ⎝ 2VT ⎠ c2 c1 tanh ( x ) = x − id a1 = 1 3 2 5 x + x − +... 3 15 αIEE ≡ Gm 2VT a3 = − αIEE 24VT3 a5 = αIEE 240VT5 EE Third-order model is accurate for v̂ id << B. Murmann a3 = 10VT a5 EE214 Winter 2010-11 – Chapter 9 26 Distortion in a BJT Differential Pair (2) A differential pair with perfectly matched transistors does not generate any even-order distortion products Any mismatch (e.g. in Is) will cause non-zero even-order terms – The resulting even order distortion products are typically smaller than the inherent odd-order distortion The HD3 performance of a BJT differential pair is better than that of a single BJT transistor HD3,BJTdiff 3 BJTdiff 1 a3 2 1 ⎛ vˆ id ⎞ ≅ vˆ id = ⎜ ⎟ 4 a1 48 ⎜⎝ VT ⎟⎠ B. Murmann 2 HD33,BJT BJT 1 ⎛ vˆ be ⎞ ≅ ⎜ ⎟ 24 ⎜⎝ VT ⎟⎠ 2 EE214 Winter 2010-11 – Chapter 9 27 Distortion in a MOS Differential Pair ⎛ V ⎞ ⎛ V ⎞ Iod = Id1 − Id2 = ISS⎜ id ⎟ 1 − ⎜ id ⎟ ⎝ VOV ⎠ ⎝ 2VOV ⎠ d2 d1 2 2 1 1 5 ⎛x⎞ x 1 − ⎜ ⎟ = x − x3 − x + ... 8 128 ⎝2⎠ id a1 = ISS ≡ Gm VO OV a3 = − ISS 3 8VOV a5 = ISS 5 128VOV SS Third-order model is accurate for v̂ id << B. Murmann EE214 Winter 2010-11 – Chapter 9 a3 = 4VOV a5 28 Feedback and Distortion Low-frequency distortion is a result of variations in the slope of an amplifier’s transfer characteristic. Feedback reduces the relative variation, and thus the distortion, to the same extent that it reduces fractional changes in gain. gain Vout Vin Note that feedback reduces distortion without reducing the output voltage range. The gain is also reduced, but additional gain can be provided with a preamplifer the operates with smaller signal swings, and therefore less distortion. B. Murmann EE214 Winter 2010-11 – Chapter 9 29 Power Series Analysis (1) + Si – Sε a – So Sfb f So = a1Sε + a 2S2ε + a3S3ε + K Sε = Si − f ⋅ So ∴ So = a1(Si − f ⋅ So ) + a 2 (Si − f ⋅ So )2 + a3 (Si − f ⋅ So )3 + K B. Murmann EE214 Winter 2010-11 – Chapter 9 30 Power Series Analysis (2) Expressing So as a power series expansion in Si So = b1Si + b 2Si2 + b3Si3 + K Substitute this expression for So into the result on the previous page and compare coefficients to find bi b1Si = a1(Si − f ⋅ b1Si ) b1 = a1 1+ a1f Thus, the feedback reduces the coefficient of the fundamental term in the forward amplifier by 1 + af B. Murmann EE214 Winter 2010-11 – Chapter 9 31 Power Series Analysis (3) Second-order terms b 2Si2 = −a1f ⋅ b 2Si2 + a 2 ((Si − f ⋅ b1Si )2 b2 = a 2 (1− b1f)2 a2 = 1+ a1f (1+ a1f)3 Third-order terms b3Si3 = −a1f ⋅ b3Si3 − 2a 2Si3 f ⋅ b 2 (1− f ⋅ b1) + a3 (Si − f ⋅ b1Si )3 b3 = a3 (1− b1f)3 − 2a 2 f ⋅ b 2 (1− f ⋅ b1) a3 (1+ a1f) − 2a 22 ⋅ f = 1+ a1f (1+ a f)5 1 B. Murmann EE214 Winter 2010-11 – Chapter 9 32 Comments Large loop gain (a1f) leads to small nonlinearity b3 contains a term due to a2 – This is due to signal interaction with the second-order term fed back to the input It is possible to obtain b3 = 0 without large loop gain B. Murmann EE214 Winter 2010-11 – Chapter 9 33 An Interesting Example VCC So = iC = a1Sε + a 2S2ε + a3S3ε + L IC = ICQ+ iC Sε = vBE = v i − f ⋅ iC Q vi ~ RE Vi f = RE a1 = gm a2 = 1 IC 2 V2 T b1 = a1 gm = 1+ a1f 1+ gmRE b3 = B. Murmann b2 = 1 IC 6 V3 T 1 IC 1 2 2 V (1+ g R )3 T m E 1 IC 1 IC (1+ g R ) − g R m E 6 V3 2 V3 m E T T (1+ gmRE )5 a3 = gmRE = 1 2 EE214 Winter 2010-11 – Chapter 9 ⇒ b3 = 0 ⇒ HD3 = 0 34 High Frequency Distortion Analysis The power series approach studied previously ignores any frequency dependence introduced by reactive elements – Sufficient for 90% of typical circuits, including some operating at RF Assuming weakly nonlinear behavior, the frequency dependence can be included using a Volterra Series model – Vito Volterra, 1887 The purpose of this handout is to provide a few basic examples that will allow you to understand the general framework Examples – Memoryless nonlinearity followed by a filter – Memoryless nonlinearity preceded and followed by a filter – RC circuit with nonlinear capacitance – RC circuit with nonlinear resistance B. Murmann EE214 Winter 2010-11 – Chapter 9 35 Example 1 ic = a1vi + a2 vi2 + a3 vi3 + ... R ic vo a1 = gm vi VI a2 = C K( jω) = 1 ICQ 2 VT2 a3 = 1 ICQ 6 VT3 vo −R = ic 1 + jωRC Ignoring device capacitances and finite output resistance for simplicity B. Murmann EE214 Winter 2010-11 – Chapter 9 36 Single-Tone Input vi = vˆ i cos ( ωt ) Ignoring DC offset and gain expansion, we have ic = a1vˆ i cos ( ωt ) + 1 1 a2 vˆ i2 cos ( 2ωt ) + a3 vˆ i3 cos ( 3ωt ) + ... 2 4 The output voltage consists of the same tones, with their magnitude and phase altered by the linear filter K(jω) v o = K ( jω) ⋅ a1vˆ i cos ( ωt + φω ) 1 + K ( 2jω) ⋅ a2 vˆ i2 cos ( 2ωt + φ2ω ) 2 φmω = ∠K ( m ⋅ jω) 1 + K ( 3jω) ⋅ a3 vˆ i3 cos ( 3ωt + φ3ω ) + ... 4 B. Murmann EE214 Winter 2010-11 – Chapter 9 37 Two-Tone Input vi = vˆ 1 cos ( ω1t ) + vˆ 2 cos ( ω2 t ) Substituting this input into the power series series, and using the identities shown below, the complete expression for the collector current is most elegantly expressed as shown on the next slide cos ( α ) cos ( β ) = cos ( α ) cos ( β ) cos ( γ ) = 1 ⎡cos ( α + β ) + cos ( α − β ) ⎤⎦ 2⎣ 1 ⎡cos ( α + β + γ ) + cos ( α + β − γ ) 4⎣ + cos ( α − β + γ ) + cos ( α − β − γ ) ⎤⎦ B. Murmann EE214 Winter 2010-11 – Chapter 9 38 Frequency Components ic = a1 ⎡⎣ vˆ 1 cos ( ω1t ) + vˆ 2 cos ( ω2 t ) ⎤⎦ + ω1, ω2 a2 ⎡ 2 vˆ 1 cos ([ ω1 ± ω1] t ) + vˆ 22 cos ([ ω2 ± ω2 ] t ) 2 ⎣ 0, 2ω1, 2ω2 +2vˆ 1 vˆ 2 cos ([ ω1 ± ω2 ] t ) ⎤ ⎦ + ω1-ω2, ω1+ω2 a3 ⎡ 3 vˆ 1 cos ([ ω1 ± ω1 ± ω1] t ) + vˆ 32 cos ([ ω2 ± ω2 ± ω2 ] t ) 4 ⎣ ω1, ω2, 3ω 3 1, 3ω 3 2 + 3vˆ 12 vˆ 2 cos ([ ω1 ± ω2 ± ω2 ] t ) ω1, 2ω1-ω2, 2ω1+ω2 + 3vˆ 1 vˆ 22 cos ([ ω1 ± ω1 ± ω2 ] t ) ⎤ + ... ⎦ ω2, 2ω2-ω1, 2ω2+ω1 B. Murmann EE214 Winter 2010-11 – Chapter 9 39 Filtered Output ( ) ( ) v o = a1 ⎡ vˆ 1 K ( jω1 ) cos ω1t + φω1 + vˆ 2 K ( jω2 ) cos ω2 t + φω2 ⎤ ⎣ ⎦ + a2 ⎡ K ( 2jω1 ) ⋅ vˆ 12 cos 2ω1t + φ2ω1 + vˆ 12 K ( 0 ) ⎣ 2 ( ) ( ) + K ( 2jω2 ) ⋅ vˆ 22 cos 2ω2 t + φ2ω2 + vˆ 22 K ( 0 ) ( ) ( ) + K ( j [ ω1 − ω2 ]) ⋅ 2vˆ 1 vˆ 2 cos [ ω1 − ω2 ] t + φω1−ω2 + K ( j [ ω1 + ω2 ]) ⋅ 2vˆ 1 vˆ 2 cos [ ω1 + ω2 ] t + φω1+ω2 ⎤ ⎦ + B. Murmann a3 [...] 4 EE214 Winter 2010-11 – Chapter 9 40 Short Hand Notation v o = a1K ( jωa ) v i + a2K ( jωa + jωb ) v i2 + a3K ( jωa + jωb + jωc ) v i3 + ... H1( jωa ) H3 ( jωa + jωb + jωc ) H2 ( jωa + jωb ) Operator “◦” means – Multiply each frequency component in vim by Hm ( jωa, jωb,...) and shift phase by ∠Hm ( jωa, jωb,...) The arguments ωa, ωb, ωc, … are auxiliary variables taking on all permutations of ω1, ±ω2, … ±ωm B. Murmann EE214 Winter 2010-11 – Chapter 9 41 General Frequency Domain Volterra Series v o = H1 ( jωa ) v i + H2 ( jωa , jωb ) v i2 + H3 ( jωa , jωb , jωc ) v i3 + ... For the circuit example discussed previously, the coefficients are given as follows H1(jωa ) = −a1R 1 + jωaRC H2 (jωa , jωb ) = −a2R 1 + ( jωa + jωb ) RC H3 (jωa , jωb , jωc ) = B. Murmann −a3R 1 + ( jωa + jωb + jωc ) RC EE214 Winter 2010-11 – Chapter 9 42 Distortion Metrics Taylor Series Volterra Series HD2 1 a2 v̂ i v 2 a1 1 H2 ( jω1, jω1 ) v̂ i v 2 H1 ( jω1 ) HD3 1 a3 2 v̂i v 4 a1 1 H3 ( jω1, jω1, jω1 ) 2 vi v̂ 4 H1 ( jω1 ) IM3 3 a3 2 v̂i 4 a1 3 H3 ( jω1, jω1, − jω2 ) 2 v̂i 4 H1 ( jω1 ) Volterra series model – Second and third order distortion still vary with square and cube of input amplitude, respectively – But, there is no fixed relationship between HD2 and IM2, and HD3 and IM3 B. Murmann EE214 Winter 2010-11 – Chapter 9 43 Distortion Metrics for Example 1 1 1 H2 ( jω, jω) 1 a2 1 + 2jωRC 1 a2 1 + jωRC HD2 = vˆ i = vˆ i = vˆ i 2 H1 ( jω) 2 a1 2 a1 1 + 2jωRC 1 1 + jωRC HD3 = IM3 = B. Murmann 1 H3 ( jω, jω, jω) 2 1 a3 1 + jωRC 2 vˆ i = vˆ i 4 4 a1 1 + 3jωRC H1 ( jω) 1 + jωRC 3 H3 ( jω1, jω1, − jω2 ) 2 1 a3 vˆ i = vˆ i2 4 4 a1 1 + j ( 2ω1 − ω2 ) RC H1 ( jω1 ) EE214 Winter 2010-11 – Chapter 9 44 HD2 and HD3 Distortion Plots for Example 1 ICQ = 1mA B. Murmann vˆ i = VT 5 EE214 Winter 2010-11 – Chapter 9 45 IM3 Distortion Plot for Example 1 ICQ = 1mA vˆ i1 = vˆ i2 = VT 5 For ω2 → ω1, the IM3 distortion product is close to the fundamental tone ω1, and therefore IM3 becomes nearly frequency independent B. Murmann EE214 Winter 2010-11 – Chapter 9 46 Example 2 2 3 ic = a1vbe + a2 vbe + a3 vbe + ... K(jω) = vo −R = ic 1 + jωRC K in (jω) = vbe 1 = vi 1 + jωRinCin Similar to example 1, but now including an additional filter at the input B. Murmann EE214 Winter 2010-11 – Chapter 9 47 Two-Tone Input vi = vˆ 1 cos ( ω1t ) + vˆ 2 cos ( ω2 t ) The two input tones are now processed by a linear filter before being sent through the nonlinearity At the base of the BJT, we have ( ) ( v be = K in ( jω1 ) ⋅ vˆ 1 cos ω1t + ψ ω1 + K in ( jω2 ) ⋅ vˆ 2 cos ω2 t + ψ ω2 ) ψmω = ∠K in ( m ⋅ jω) In short hand notation, this can be written as v be = K in ( jωa ) v i B. Murmann EE214 Winter 2010-11 – Chapter 9 48 2 3 v o = a1K ( jωa ) v be + a2K ( jωa + jωb ) v be + a3K ( jωa + jωb + jωc ) v be + ... v o = a1K ( jωa ) ⎡⎣K in ( jωa ) v i ⎤⎦ + a2K ( jωa + jωb ) ⎡⎣K in ( jωa ) v i ⎦⎤ 2 + a3K ( jωa + jωb + jωc ) ⎡⎣K in ( jωa ) v i ⎤⎦ 3 First order term a1K ( jωa ) ⎡⎣K in ( jωa ) v i ⎤⎦ = a1K ( jωa ) K in ( jωa ) v i B. Murmann EE214 Winter 2010-11 – Chapter 9 49 Second order term 2 a2K ( jωa + jωb ) ⎡⎣K in ( jωa ) v i ⎤⎦ = ? 2 ⎡⎣K in ( jωa ) v i ⎤⎦ = ⎡⎣K in ( jωa ) {vˆ 1 cos ( ω1t ) + vˆ 2 cos ( ω2t )}⎤⎦ ( 2 ) ( ) = ⎡ K in ( jω1 ) ⋅ vˆ 1 cos ω1t + ψ ω1 + K in ( jω2 ) ⋅ vˆ 2 cos ω2 t + ψ ω2 ⎤ ⎣ ⎦ ( 2 = K in ( jω1 ) ⋅ vˆ 12 cos {ω1 ± ω1} t + ψ ω1 ± ψ ω1 2 ( ) + K in ( jω2 ) ⋅ vˆ 22 cos {ω2 ± ω2 } t + ψ ω2 ± ψ ω2 ( ) ˆ 1 vˆ 2 cos {ω1 ± ω2 } t + ψ ω ± ψ ω + K in i ( jω1 ) K iin ( jω2 ) ⋅ v 1 2 ) = K in ( jωa ) K in ( jωb ) v i2 B. Murmann EE214 Winter 2010-11 – Chapter 9 50 2 Coefficients H1( jωa ) = −a1R 1 1 + jωaRC 1 + jωaRC H2 ( jωa , jωb ) = −a2R 1 1 1 + jωaRC 1 + jωbRC 1 + ( jωa + jωb ) RC H3 ( jωa , jωb , jωc ) = −a3R 1 1 1 1 + jωaRC 1 + jωbRC 1 + jωcRC 1 + ( jωa + jωb + jωc ) RC B. Murmann EE214 Winter 2010-11 – Chapter 9 51 Example 3 i ij [[Chun & Murmann,, JSSC 10/2006]] Cj models the nonlinear capacitance of an electrostatic discharge (ESD) protection device (e.g. a basic diode structure as shown to the right) B. Murmann EE214 Winter 2010-11 – Chapter 9 52 Junction Capacitance Model Cj = Using C j0 = M ⎛ VOQ + v o ⎞ ⎜1+ ⎟ ψ0 ⎝ ⎠ C jQ = C j0 1 M C j0 1 M (1 + x ) VR = VOQ + ψ0 b1 = − Cj = C jQ M ⎛ vo ⎞ ⎜1+ ⎟ ⎝ VR ⎠ B. Murmann = 1 − Mx + M ⎛ VOQ ⎞ ⎜1+ ⎟ ψ0 ⎠ ⎝ we can write M ⎛ ψ0 + VOQ ⎞ ⎛ ψ0 + VOQ + v o ⎞ ⎜ ⎟ ⎜ ⎟ ψ0 ⎝ ⎠ ⎝ ψ0 + VOQ Q ⎠ M VR b2 = ( ) 1 M + M2 x 2 + ... 2 M + M2 2VR = C jQ ⎡1 + b1v o + b2 v o2 + ...⎤ ⎣ ⎦ EE214 Winter 2010-11 – Chapter 9 53 Circuit Analysis ij = Cj ⎡ dv dv o dv 1 dv 2 1 dv 3 ⎤ = C jQ ⎡1 + b1v o + b2v o2 + ...⎤ o = C jQ ⎢ o + b1 o + b3 o ⎥ ⎣ ⎦ dt dt dt 3 dt ⎥⎦ ⎢⎣ dt 2 ( i = C jQ + CI ⎡ ⎣⎢ ( vi = v o + i ⋅ R = v o + R C jQ + CI vi = v o + RC0 C0 = C jQ + CI B. Murmann 2 3⎤ ) dvdto + C jQ ⎢ 21 b1 dvdto + 31 b3 dvdto ⎥ ) ⎦⎥ ⎡ 1 dv 2 1 dv 3 ⎤ dv o + RC jQ ⎢ b1 o + b3 o ⎥ dt dt 3 dt ⎥⎦ ⎢⎣ 2 d d d v o + RC1 v o2 + RC2 v o3 dt dt dt MC jQ b C1 = 1 C jQ = − 2 2VR ( ) M + M2 C jQ b1 C2 = C jQ = − 3 6VR2 EE214 Winter 2010-11 – Chapter 9 54 Volterra Series We W are looking l ki ffor a Volterra V lt series i representation t ti off the th form f v o = H1 ( jωa ) v i + H2 ( jωa , jωb ) v i2 + H3 ( jωa , jωb , jωc ) v i3 + ... The coefficients H1, H2 and H3 can be found by inserting the above series into the nonlinear differential equation shown on the previous slide and subsequently comparing the coefficients on the LHS and RHS of the equation vi = H1 ( jωa ) vi + H2 ( jωa , jωb ) vi2 + H3 ( jωa , jωb , jωc ) vi3 + RC0 d⎡ H1 ( jωa ) vi + H2 ( jωa , jωb ) vi2 + H3 ( jωa , jωb , jωc ) vi3 ⎤ ⎦ dt ⎣ + RC1 2 d⎡ H1 ( jωa ) vi + H2 ( jωa , jωb ) vi2 + H3 ( jωa , jωb , jωc ) vi3 ⎤ ⎦ dt ⎣ + RC2 3 d⎡ H1 ( jωa ) vi + H2 ( jωa , jωb ) vi2 + H3 ( jωa , jωb , jωc ) vi3 ⎤ ⎦ dt ⎣ B. Murmann EE214 Winter 2010-11 – Chapter 9 55 Coefficient Comparison (1) First order d⎤ ⎡ 1 vi = ⎢1 + RC0 ⎥ H1 ( jωa ) vi dt ⎦ ⎣ d⎤ ⎡ 1 = ⎢1 + RC0 ⎥ H1 ( jωa ) = ⎡⎣1 + RC0 jωa ⎤⎦ H1 ( jωa ) dt ⎦ ⎣ ∴ H1 ( jωa ) = 1 1 + RC0 jωa This is just the linear transfer function as expected B. Murmann EE214 Winter 2010-11 – Chapter 9 56 Coefficient Comparison (2) Second order 2 d⎤ d ⎡ 0 vi2 = ⎢1 + RC0 ⎥ H2 ( jωa , jωb ) vi2 + RC1 ⎡H1 ( jωa ) vi ⎤ ⎦ dt ⎦ dt ⎣ ⎣ Simplify using the following rules d H2 ( jωa , jωb ) = ( jωa + jωb ) H2 ( jωa , jωb ) dt 2 ⎡H1 ( jωa ) vi ⎤ = H1 ( jωa ) H1 ( jωb ) vi2 ⎣ ⎦ d ⎡H1 ( jωa ) H1 ( jωb ) ⎤⎦ = ( jωa + jωb ) H1 ( jωa ) H1 ( jωb ) dt ⎣ B. Murmann EE214 Winter 2010-11 – Chapter 9 57 Coefficient Comparison (3) 0 = ⎡⎣1 + RC0 ( jωa + jωb ) ⎤⎦ H2 ( jωa , jωb ) + RC1 ( jωa + jωb ) H1 ( jωa ) H1 ( jωb ) H2 ( jωa , jωb ) = −RC1 ( jωa + jωb ) H1 ( jωa ) H1 ( jωb ) 1 + RC0 ( jωa + jωb ) ∴ H2 ( jωa , jωb ) = −RC1 ( jωa + jωb ) H1 ( jωa ) H1 ( jωb ) H1 ( jωa + jωb ) H2 becomes zero for f ω→0; this makes intuitive sense B. Murmann EE214 Winter 2010-11 – Chapter 9 58 Coefficient Comparison (4) Third order d⎤ ⎡ 0 vi3 = ⎢1 + RC0 ⎥ H3 ( jωa , jωb , jωc ) vi3 dt ⎦ ⎣ ( + RC1 d⎡ 2 H1 ( jωa ) vi dt ⎣ + RC2 3 d⎡ H1 ( jωa ) vi ⎤ ⎦ dt ⎣ ) (H ( jω , jω ) v )⎤⎦ 2 a b 2 i Second-order Interaction Term Simplifying and neglecting the second-order interaction term yields ∴ H3 ( jωa , jωb , jωc ) = −RC2 ( jωa + jωb + jωc ) H1 ( jωa ) H1 ( jωb ) H1 ( jωc ) ⋅ H1 ( jωa + jωb + jωc ) B. Murmann EE214 Winter 2010-11 – Chapter 9 59 Harmonic Distortion Expressions (1) 2 RC1 ( 2jω) ⎡⎣H1 ( jω) ⎤⎦ H1 ( 2jω) 1 H2 ( jω, jω) 1 HD2 = vˆ i = vˆ i = ωRC1 H1 ( jω) H1 ( 2jω) vˆ i 2 H1 ( jω) 2 H1 ( jω) ∴ HD2 = ⎛ v̂ ⎞ 1 MωRC jQ H1 ( jω) H1 ( 2jω) ⎜ i ⎟ ⎜ VR ⎟ 2 ⎝ ⎠ HD2 improves for – Lower swing (vi/VR) – Lower frequency – Lower drive resistance (R) – Smaller capacitive nonlinearity (M→0) and smaller CjQ Adding extra linear capacitance (CI) will lower the corner frequency of H1 – Assuming that we cannot tolerate much signal attenuation, this won’t help reduce the distortion all that much in the useable frequency range B. Murmann EE214 Winter 2010-11 – Chapter 9 60 Harmonic Distortion Expressions (2) RC2 ( 3jω) (H1 ( jω) ) H1 ( 3jω) 1 H3 ( jω, jω, jω) 2 1 HD3 = vˆ i = vˆ i2 4 4 H1 ( jω) H1 ( jω) 3 = 2 3 ωRC2 H1 ( jω) H1 ( 3jω) vˆ i2 4 ⎛ v̂ ⎞ 1 ∴ HD3 = M + M2 ωRC jQ H1 ( jω) H1 ( 3jω) ⎜ i ⎟ ⎜ VR ⎟ 8 ⎝ ⎠ ( ) 2 HD3 behaves similar to HD2 – The distortion depends on similar quantities and cannot be improved by adding additional linear capacitance (CI) HD3 is often more critical than HD2, since the latter can be improved significantly by employing a differential circuit topology B. Murmann EE214 Winter 2010-11 – Chapter 9 61 Plot v̂ i = 0.5V v 0 5V R = 25Ω CI = 0 C jQ = 1pF M = 0.3 03 VR = VOQ + ψ0 = 2.2V B. Murmann EE214 Winter 2010-11 – Chapter 9 62 Example 4 An A RC circuit i it with ith a lilinear capacitor, it b butt nonlinear li resistor i t The circuit below implements a track-and-hold circuit used in switched capacitor circuits (filters, A/D converters, etc.) – More in EE315A,B The MOSFET is used as a switch and operates in the triode region, exhibiting nonlinear resistive behavior [W. Yu, IEEE TCAS II, 2/1999] K 2 ID ≅ K ( VGS − Vt ) VDS − VDS 2 B. Murmann EE214 Winter 2010-11 – Chapter 9 63 Analysis (1) ID ≅ K ( VGS − Vt ) VDS − C K 2 VDS 2 dVo K 2 = K ( ϕ − Vo − Vt )( Vi − Vo ) − ( Vi − Vo ) dt 2 Solving for the coefficients of the Volterra series linking vo and vi, and evaluating HD2 and HD3 yields HD2 = ⎛ ⎞ v̂ i 1 ωRC ⎜ ⎟ 2 ⎝ VGS − Vt ⎠ ⎛ ⎞ v̂ i 1 HD3 = ωRC ⎜ ⎟ 4 ⎝ VGS − Vt ⎠ 2 where VGS is the quiescent point gate-source voltage and R is the quiescent point resistance of the MOSFET, i.e. K[VGS-Vt]-1 B. Murmann EE214 Winter 2010-11 – Chapter 9 64 Analysis (2) For low distortion – Make signal amplitude much smaller than VGS-Vt • Low swing, which is usually undesired – Make 1/RC much larger than 2π·fin • Small C or big MOSFET, which is usually undesired Example HD3 calculation ⎫ ⎪ ⎪ ⎪ vˆ i = 0.2V 0 2V ⎪ ⎪⎪ V VOQ = VIQ = DD = 0.9V ⎬ 2 ⎪ ⎪ VGS − Vt = VDD − VOQ − Vt ⎪ ⎪ = 1.8V − 0.9V − 0.4V = 0.45V ⎪ ⎪⎭ ω= B. Murmann 1 1 10 RC 2 HD3 = EE214 Winter 2010-11 – Chapter 9 1 1 ⎛ 0.2 ⎞ = −46dB 4 10 ⎜⎝ 0.45 ⎟⎠ 65 Example 5 [Terrovitis & Meyer, JSSC 10/2000] B. Murmann EE214 Winter 2010-11 – Chapter 9 66 B. Murmann EE214 Winter 2010-11 – Chapter 9 67 Additional Topics in Distortion Analysis Covered in EE314 – RF-centric metrics • Intercept points • 1-dB gain compression point Other – Nonlinearities in passive components – Distortion cancelation techniques – Cascading nonlinearities – Series reversion – Distortion in clipped or amplitude limiting waveforms • E.g. in oscillators B. Murmann EE214 Winter 2010-11 – Chapter 9 68 References (1) D. O. Pederson and K. Mayaram, Analog Integrated Circuits for Communication, Springer, 1991 P. Wambacq and W. M. C. Sansen, Distortion Analysis of Analog Integrated Circuits, Springer, 1998 Hurst, Lewis, Gray y & Meyer, y 5th edition, pp pp. 355-359 – Distortion analysis of a source follower B. Razavi, Design of Analog CMOS Integrated Circuits, McGraw-Hill, 2000, Chapter 13 W. Sansen, “Distortion in elementary transistor circuits,” IEEE Trans. Circuits and Systems II, vol. 46, pp. 315-325, March 1999 P P. Wambacq, Wambacq G. G Gielen, Gielen and P P. Kinget, Kinget “High-Frequency “High Freq enc Distortion Analysis of Analog Integrated Circuits,” IEEE Trans. Circuits and Syst. II, pp. 335-345, March 1999 B. Murmann EE214 Winter 2010-11 – Chapter 9 69 References (2) W W. Rugh, R h Nonlinear N li S System t Th Theory, http://rfic.eecs.berkeley.edu/ee242/pdf/volterra_book.pdf R. G. Meyer and M. L. Stephens, "Distortion in variable-capacitance di d " IEEE J. diodes," J Solid-State S lid St t Circuits, Ci it pp. 47-54, 47 54 F Feb. b 1975 K. L. Fong and R.G. Meyer, "High-frequency nonlinearity analysis of common-emitter and differential-pair transconductance stages," IEEE J. Solid State Circuits, Solid-State Circuits pp pp.548-555, 548 555 Apr Apr. 1998 M. T. Terrovitis and R. G. Meyer, "Intermodulation distortion in currentcommutating CMOS mixers," IEEE J. Solid-State Circuits, pp.14611473 Oct 1473, Oct. 2000 J. Chun and B. Murmann, "Analysis and Measurement of Signal Distortion due to ESD Protection Circuits," IEEE J. Solid-State Circuits, pp 2354-2358 pp. 2354-2358, Oct Oct. 2006 W. Yu, S. Sen and B. H. Leung, "Distortion analysis of MOS track-andhold sampling mixers using time-varying Volterra series," IEEE Trans. Circuits and Syst Syst. II II, pp pp. 101-113 101 113, Feb Feb. 1999 B. Murmann EE214 Winter 2010-11 – Chapter 9 70 Ch t 10 Chapter Operational Amplifiers and Output Stages B. Murmann Stanford University Reading Material: Sections 5.2, 5.5, 6.8, 7.4, 9.4.5 http://www nxp com/documents/data sheet/NE5234 SA5234 pdf http://www.nxp.com/documents/data_sheet/NE5234_SA5234.pdf B. Murmann EE214 Winter 2010-11 – Chapter 10 2 B. Murmann EE214 Winter 2010-11 – Chapter 10 3 Architecture Provides differential input with large common mode d range Provides additional voltage lt gain i Provides some g g gain,, voltage approximately independent of common mode B. Murmann EE214 Winter 2010-11 – Chapter 10 Provides high current drive capability bilit Output can (approximately) g rail-to-rail swing 4 Basic PNP Differential Pair Input Stage Circuit works for input common mode down to VEE (0V), as long as VBE > VCE(sat ) + VR1 VR1 < VBE − VCE(sat ) ≅ 0.8V 0 8V − 0 0.2V 2V = 0 0.6V 6V This limits the gain we can extract from this stage gm1R1 = IBIAS V 600mV R1 = R1 = = 23 2VT VT 26mV 26 V Text, p. 449 B. Murmann EE214 Winter 2010-11 – Chapter 10 5 PNP Folded Cascode Input Stage Circuit still works for input common mode down to VEE (0V), as before But, the low frequency voltage gain is much larger g g v od R3 R = gm1 1 o v id R3 + gm Ro ≅ ro6 (1 + gm6R6 ) ro3 (1 + gm3R3 ) B. Murmann EE214 Winter 2010-11 – Chapter 10 6 NE5234 Input Stage X ~0.9V Provides rail-to rail common-mode input range by combining the currents from a PNP and NPN pair The NPN pair “steals” current from the PNP pair at node X whenever the input common mode is greater than |VBE| – Total Gm is approximately constant B. Murmann EE214 Winter 2010-11 – Chapter 10 7 Second Stage 0.5V 0.5V 0.5V ~0.9V + 0.4V - B. Murmann EE214 Winter 2010-11 – Chapter 10 8 Common Mode Operating Point of Stage 1 Output B. Murmann EE214 Winter 2010-11 – Chapter 10 9 Basic Emitter Follower (“Class-A”) Output Stage Issues – Output p voltage g cannot g go higher than VCC – VBE – Need large quiescent current IQ to provide good current sinking capability – IQ flows even if no signal is present, i.e. Io=0 B. Murmann EE214 Winter 2010-11 – Chapter 10 10 Output Stage Nomenclature Cl Class-A A – Output devices conduct for entire cycle of output sine wave Class-B – Output devices conduct for ≅50% of sine wave cycle Class-AB – Output p devices conduct for >50%,, but <100% of cycle y Class-A B. Murmann Class-B Class-AB EE214 Winter 2010-11 – Chapter 10 11 Class-AB Emitter Follower Output Stage Small quiescent current, but large drive capability Remaining issue – Output voltage swing is severely limited B. Murmann EE214 Winter 2010-11 – Chapter 10 12 Class-AB Common Emitter Output Stage Q1 and Q2 are large devices Want to set the quiescent point such that IC1 and IC2 are small when no signal is applied B. Murmann EE214 Winter 2010-11 – Chapter 10 13 Analysis Assuming IC3 = IC4 = IREF by symmetry, we can write VBE6 + VBE8 = VBE4 + VBE2 IREF IREF IREF IC2 = Is6 Is8 Is4 Is2 IC2 = IREF Is2 Is4 Is6 Is8 Quiescent current is set by y IREF and emitter area ratios ((Is ∝ AE) The same principle applies to CMOS implementations – Original paper by Monticelli, JSSC 12/1986 B. Murmann EE214 Winter 2010-11 – Chapter 10 14 CMOS Version [Hogervorst, JSSC 12/1994] B. Murmann EE214 Winter 2010-11 – Chapter 10 15 Simulated Currents (Vo=0) Current [A] Iout Output transistors never turn off Quiescent current set by transistor ratios Large drive capability ID(M26) ID(M25) Iin1=Iin2 [A] B. Murmann EE214 Winter 2010-11 – Chapter 10 16 NE 5234 Output Stage Feedback through Q39-40 ensures that both transistors remain on at extreme output swings B. Murmann EE214 Winter 2010-11 – Chapter 10 17 Simplified AC Schematic of Complete OpAmp Nested Miller Compensation B. Murmann EE214 Winter 2010-11 – Chapter 10 18 Compensation Issue (1) The designer of a general purpose OpAmp knows nothing about the feedback network that the user will connect around the amplifier Therefore, general purpose OpAmps are typically compensated for the "worst case", i.e. unity feedback configuration This tends to be wasteful, since much less compensation p may y be needed for other feedback configurations – To overcome this issue, some general purpose OpAmps provide an external pin to let the user decide on the required compensation capacitor it B. Murmann EE214 Winter 2010-11 – Chapter 10 19 Compensation Issue (2) Wasted BW B. Murmann EE214 Winter 2010-11 – Chapter 10 20 Slewing P Previous i analyses l h have been b concerned d with ith th the small-signal ll i lb behavior h i of feedback amplifiers at high frequencies Sometimes the behavior with large input signals (either step inputs or sinusoidal i id l signals) i l ) iis also l off interest i t t – See e.g. switched capacitor circuits in EE315A Example using NE5234 B. Murmann EE214 Winter 2010-11 – Chapter 10 21 Output Response – Expected versus Actual B. Murmann EE214 Winter 2010-11 – Chapter 10 22 Investigation B. Murmann EE214 Winter 2010-11 – Chapter 10 23 General Analysis Slew Rate B. Murmann EE214 Winter 2010-11 – Chapter 10 24 “Internal” Amplifiers (1) Most amplifiers used in large systems-on-chip (SoC) are not as complex as a typical general purpose OpAmps – Typically, no output stage is needed, since the loads are either small and/or purely capacitive Operational Transconductance Amplifier (OTA) = OpAmp minus Output Stage More in EE315A (see also text, section 6.1.7, and chapter 12) B. Murmann Mehta et al., "A 1.9GHz Single-Chip CMOS PHS Cellphone," ISSCC 2006. EE214 Winter 2010-11 – Chapter 10 25 “Internal” Amplifiers (2) B. Murmann EE214 Winter 2010-11 – Chapter 10 26

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